Sense amplifier for bidirectional sensing of memory cells of a non-volatile memory

ABSTRACT

A sense amplifier for a memory circuit is presented that can sense a selected memory cell in either a first sensing mode, in which current from the selected memory cell flows from the memory cells into the sense amplifier, or a second sensing mode, in which current is discharged from the sense amplifier through the selected memory cell. In the first sensing mode, current from a selected memory cell is conducted through cascaded PMOS transistors to charge a sensing node, with the resultant voltage level on the sensing node used to determine the result of the sensing operation.

BACKGROUND

Semiconductor memory is widely used in various electronic devices such as cellular telephones, digital cameras, personal digital assistants, medical electronics, mobile computing devices, servers, solid state drives, non-mobile computing devices and other devices. Semiconductor memory may comprise non-volatile memory or volatile memory. A non-volatile memory allows information to be stored and retained even when the non-volatile memory is not connected to a source of power (e.g., a battery).

BRIEF DESCRIPTION OF THE DRAWINGS

Like-numbered elements refer to common components in the different figures.

FIG. 1A is a functional block diagram of a memory device.

FIG. 1B is a block diagram depicting one example of a memory structure.

FIG. 2 is a block diagram depicting one embodiment of a memory system.

FIG. 3 is a perspective view of a portion of one embodiment of a monolithic three-dimensional memory structure.

FIG. 4 is a schematic of a plurality of NAND strings.

FIG. 5 depicts various embodiments of a portion of a three-dimensional memory array with a vertical cross-point structure.

FIG. 6 depicts threshold voltage distributions in a three bit per cell embodiment.

FIG. 7A is a flowchart describing one embodiment of a process for programming/writing.

FIG. 7B is a flowchart describing one embodiment of a process for programming/writing data into memory cells connected to a common word line.

FIG. 7C depicts a word line voltage during programming/writing and verify operations.

FIG. 8 is a flowchart describing one embodiment of a process for reading data from non-volatile memory cells.

FIGS. 9A and 9B illustrate a window of threshold values to which memory cells can be programmed, including negative threshold values.

FIGS. 10A-10C illustrate several techniques for sensing memory cells with negative threshold voltage values.

FIG. 11 illustrates some elements of an embodiment of a sense amplifier that can be used to perform a sensing operation using the techniques illustrated in FIG. 10C.

FIG. 12 is a more detailed representation of an embodiment for the sense amplifier of FIG. 11.

FIG. 13 illustrates one embodiment for some of the control waveforms for a sensing operation using the embodiment of FIG. 12.

FIG. 14 is a flowchart describing one embodiment of a process for a sensing operation for the sense amplifier of FIG. 12 using the waveforms of FIG. 13.

FIG. 15 is a circuit diagram of an additional embodiment of a sense amplifier operable in a first mode and a second mode.

FIG. 16 is a flowchart presenting an embodiment for a sensing operation in which the sensing node in the sense amplifier circuit of FIG. 15 is discharged through a selected memory cell.

FIG. 17 illustrates an embodiment for the discharge of a sensing node in the sense amplifier circuit of FIG. 15 through a selected memory cell.

FIG. 18 illustrates a pre-charge phase for a sensing operation of the embodiment of FIG. 15 in a sensing mode where the current from a selected memory cell flows into the sense amplifier.

FIG. 19 illustrates a settling phase, in which the voltage levels on the sense amplifier circuit are stabilized for a sensing operation of the embodiment of FIG. 15 in a sensing mode where the current from a selected memory cell flows into the sense amplifier.

FIG. 20 illustrates the charging of the sensing node charging phase for a sensing operation of the embodiment of FIG. 15 in a sensing mode where the current from a selected memory cell flows into the sense amplifier.

FIG. 21 illustrates the strobe phase for a sensing operation of the embodiment of FIG. 15 in a sensing mode where the current from a selected memory cell flows into the sense amplifier.

FIG. 22 is a circuit diagram of an alternate embodiment of a sense amplifier operable in a first mode and a second mode.

FIG. 23 is a diagram of an embodiment of a bias-voltage generation circuit that can provide control gate voltages for some of the PMOS elements of the sense amplifier embodiments in FIGS. 15 and 22.

FIG. 24 is a diagram of an embodiment of a bias-voltage generation circuit that can provide the bias level used in discharging the sensing node of the sense amplifier embodiments in FIGS. 15 and 22.

FIG. 25 is a flowchart describing one embodiment of a process for a sensing operation for the sense amplifier of FIG. 15 in the first sensing mode, in which current flows into the sense amplifier.

DETAILED DESCRIPTION

To increase the amount of data stored on a non-volatile memory device, data can be stored in a multi-level cell (MLC) format, where an individual memory cell can be programmed to multiple different states, where each memory cell can hold more than one bit of data. In memory cells where different data states correspond to different threshold voltage (Vt) values, this involves splitting up the range, or window, of available Vt values into a number of ranges corresponding to the different data states. To store more states per cell, the Vt range allotted to each state needs to be made smaller, the size of the window increased, or both. The size of the Vt window can be increased by extending the window further into negative Vt values and having multiple states with negative, or non-positive, Vt values. However, for this to be useful, the memory device must be able to distinguish between different non-positive Vt states.

Sensing negative Vt states by most standard sensing techniques and sense amplifier structures has a number of limitations. In a typical sensing arrangement, the control gate of a memory cell is biased by a read voltage and a bit line connected to sense amplifier is discharged through the memory cell to a source line, where the amount of discharge depends on the value of the read voltage relative to the memory cell's Vt. Under this usual arrangement, reading of negative Vt states uses negative read voltages; however, negative voltages are typically not available on a memory die and their introduction involves complications. Alternately, negative Vt states can be read by raising the source voltage for the memory cell, but this approach can usually only extend to a fairly shallow negative Vt range. To allow for sensing more deeply into the negative Vt range, the following introduces sense amplifier structures and techniques in which the source is discharged through a selected memory cell into the bit line and sense amplifier, reversing the usual direction of current flow through the selected memory cell in a sensing operation.

More specifically, a first set of embodiments for a sense amplifier structure and sensing techniques are described where, in a first phase, the source line is discharged through a selected memory to the corresponding bit line and on into the sense amp. The amount of current discharged in this phase will depend on the conductivity of the memory cell, which in turn depends on the word line voltage supplied to the control gate of the selected memory cell relative to its threshold voltage. A discharge transistor has its control gate connected to the memory cell's discharge path during the first phase, so that the conductivity of the discharge transistor will reflect the conductivity of the selected memory cell. The control gate of the discharge transistor is then set to float at this level. In a second phase, a sense node is then discharged through the discharge transistor: as the conductivity of the discharge transistor reflects the conductivity of the selected memory cell, the rate at which the sense node discharges reflects the conductivity of the memory cell. After discharging the sense node for a sensing period, the level on the sense node is latched for the read result.

To improve accuracy of the sensing operation for the first set of embodiments, elements can be included in the sense amp to reduce noise levels. To reduce noise on the control gate of the discharge transistor when transitioning between phases, a decoupling capacitor can be connected to the control gate. The capacitor can also be biased to adjust for operating conditions, such as temperature, and device processing variations. To reduce noise on the source node of the discharge transistor, an auxiliary keeper current can be supplied through the discharge transistor during the transition between phases and on into the sense node discharge phase.

In another set of embodiments for a sense amplifier, the sense amplifier is operable in a first sensing mode, in which current flows from a selected memory through its corresponding bit line and into the sense amplifier, with the charge from the current accumulating on a sensing node and capacitor. The amount of charge accumulated on the sensing node depends on the state of the selected memory cell. In a second sensing mode, charge is discharged from the sensing node and sensing capacitor through the selected memory cell, where the amount of charge remaining on sensing node depends on the state of the selected memory cell. In either sensing mode, the state of the selected memory is then determined based on the amount of charge on the sensing node after the charging (in the first sensing mode) or discharging (in the second sensing mode) of the sensing node. In the second sensing mode, the sensing node discharges to the selected memory cell's bit line through a path of series connected NMOS transistors. For the first sensing mode, a current path from the selected memory cell to the sensing node includes cascaded PMOS transistors, where the cascaded PMOS transistors are connected in parallel with the series connected NMOS transistors used for the second sensing mode.

FIGS. 1A-5 describe examples of memory systems that can be used to implement the technology proposed herein. FIG. 1A is a functional block diagram of an example memory system 100. In one embodiment, the components depicted in FIG. 1A are electrical circuits. Memory system 100 includes one or more memory dies 108. The one or more memory dies 108 can be complete memory dies or partial memory dies. In one embodiment, each memory die 108 includes a memory structure 126, control circuitry 110, and read/write circuits 128. Memory structure 126 is addressable by word lines via a row decoder 124 and by bit lines via a column decoder 132. The row decoder 124 can include the drivers and other elements to bias the word lines for the different memory operations. The read/write circuits 128 include multiple sense blocks 150 including SB1, SB2, . . . , SBp (sensing circuitry) and allow a page of memory cells to be read or programmed in parallel, where a page is the unit in which data is written and/or read. A physical page is the physical unit of a number of cells into which data can be concurrently written and/or read, and a logical page a corresponding logical unit of data written into a physical page. More detail on sense amplifier circuits that can be used in the sense blocks 150 including SB1, SB2, . . . , SBp is given below with respect to FIGS. 11-14.

In some systems, a controller 122 is included in the same package (e.g., a removable storage card) as the one or more memory die 108. However, in other systems, the controller can be separated from the memory die 108. In some embodiments the controller will be on a different die than the memory die 108. In some embodiments, one controller 122 will communicate with multiple memory die 108. In other embodiments, each memory die 108 has its own controller. Commands and data are transferred between a host 140 and controller 122 via a data bus 120, and between controller 122 and the one or more memory die 108 via lines 118. In one embodiment, memory die 108 includes a set of input and/or output (I/O) pins that connect to lines 118.

Control circuitry 110 cooperates with the read/write circuits 128 to perform memory operations (e.g., write, read, and others) on memory structure 126, and includes a state machine 112, an on-chip address decoder 114, and a power control circuit 116. The state machine 112 provides die-level control of memory operations. In one embodiment, state machine 112 is programmable by software. In other embodiments, state machine 112 does not use software and is completely implemented in hardware (e.g., electrical circuits). In other embodiments, state machine 112 can be replaced by a programmable microcontroller. Control circuitry 110 also includes buffers such as registers, ROM fuses and other storage devices for storing default values such as base voltages and other parameters.

The on-chip address decoder 114 provides an address interface between addresses used by host 140 or controller 122 to the hardware address used by the decoders 124 and 132. Power control module 116 controls the power and voltages supplied to the word lines and bit lines during memory operations. Power control module 116 may include charge pumps for creating voltages. The sense blocks include bit line drivers.

State machine 112 and/or controller 122 (or equivalently functioned circuits), in combination with all or a subset of the other circuits depicted in FIG. 2, can be considered a control circuit that performs the functions described herein. The control circuit can include hardware only or a combination of hardware and software (including firmware). For example, a controller programmed by firmware to perform the functions described herein is one example of a control circuit. A control circuit can include a processor, FGA, ASIC, integrated circuit or other type of circuit.

The (on-chip or off-chip) controller 122 (which in one embodiment is an electrical circuit) may comprise one or more processors 122 c, ROM 122 a, RAM 122 b, a memory interface (MI) 122 d and a host interface (HI) 122 e, all of which are interconnected. The storage devices (ROM 122 a, RAM 122 b) store code (software) such as a set of instructions (including firmware), and one or more processors 122 c is/are operable to execute the set of instructions to provide the functionality described herein. Alternatively, or additionally, one or more processors 122 c can access code from a storage device in the memory structure, such as a reserved area of memory cells connected to one or more word lines. RAM 122 b can be to store data for controller 122, including caching program data (discussed below). Memory interface 122 d, in communication with ROM 122 a, RAM 122 b and processor 122 c, is an electrical circuit that provides an electrical interface between controller 122 and one or more memory die 108. For example, memory interface 122 d can change the format or timing of signals, provide a buffer, isolate from surges, latch I/O, etc. One or more processors 122 c can issue commands to control circuitry 110 (or another component of memory die 108) via Memory Interface 122 d. Host interface 122 e provides an electrical interface with host 140 data bus 120 in order to receive commands, addresses and/or data from host 140 to provide data and/or status to host 140.

In one embodiment, memory structure 126 comprises a three-dimensional memory array of non-volatile memory cells in which multiple memory levels are formed above a single substrate, such as a wafer. The memory structure may comprise any type of non-volatile memory that are monolithically formed in one or more physical levels of arrays of memory cells having an active area disposed above a silicon (or other type of) substrate. In one example, the non-volatile memory cells comprise vertical NAND strings with charge-trapping material such as described, for example, in U.S. Pat. No. 9,721,662, incorporated herein by reference in its entirety.

In another embodiment, memory structure 126 comprises a two-dimensional memory array of non-volatile memory cells. In one example, the non-volatile memory cells are NAND flash memory cells utilizing floating gates such as described, for example, in U.S. Pat. No. 9,082,502, incorporated herein by reference in its entirety. Other types of memory cells (e.g., NOR-type flash memory) can also be used.

The exact type of memory array architecture or memory cell included in memory structure 126 is not limited to the examples above. Many different types of memory array architectures or memory technologies can be used to form memory structure 126. No particular non-volatile memory technology is required for purposes of the new claimed embodiments proposed herein. Other examples of suitable technologies for memory cells of the memory structure 126 include ReRAM memories, magnetoresistive memory (e.g., MRAM, Spin Transfer Torque MRAM, Spin Orbit Torque MRAM), phase change memory (e.g., PCM), and the like. Examples of suitable technologies for memory cell architectures of the memory structure 126 include two dimensional arrays, three dimensional arrays, cross-point arrays, stacked two dimensional arrays, vertical bit line arrays, and the like.

One example of a ReRAM, or PCMRAM, cross point memory includes reversible resistance-switching elements arranged in cross point arrays accessed by X lines and Y lines (e.g., word lines and bit lines). In another embodiment, the memory cells may include conductive bridge memory elements. A conductive bridge memory element may also be referred to as a programmable metallization cell. A conductive bridge memory element may be used as a state change element based on the physical relocation of ions within a solid electrolyte. In some cases, a conductive bridge memory element may include two solid metal electrodes, one relatively inert (e.g., tungsten) and the other electrochemically active (e.g., silver or copper), with a thin film of the solid electrolyte between the two electrodes. As temperature increases, the mobility of the ions also increases causing the programming threshold for the conductive bridge memory cell to decrease. Thus, the conductive bridge memory element may have a wide range of programming thresholds over temperature.

Magnetoresistive memory (MRAM) stores data by magnetic storage elements. The elements are formed from two ferromagnetic plates, each of which can hold a magnetization, separated by a thin insulating layer. One of the two plates is a permanent magnet set to a particular polarity; the other plate's magnetization can be changed to match that of an external field to store memory. A memory device is built from a grid of such memory cells. In one embodiment for programming, each memory cell lies between a pair of write lines arranged at right angles to each other, parallel to the cell, one above and one below the cell. When current is passed through them, an induced magnetic field is created.

Phase change memory (PCM) exploits the unique behavior of chalcogenide glass. One embodiment uses a GeTe—Sb2Te3 super lattice to achieve non-thermal phase changes by simply changing the co-ordination state of the Germanium atoms with a laser pulse (or light pulse from another source). Therefore, the doses of programming are laser pulses. The memory cells can be inhibited by blocking the memory cells from receiving the light. In other embodiments, the memory cells of a PCM memory can have their data state set or reset through use of current pulses. Note that the use of “pulse” in this document does not require a square pulse, but includes a (continuous or non-continuous) vibration or burst of sound, current, voltage light, or other wave.

A person of ordinary skill in the art will recognize that the technology described herein is not limited to a single specific memory structure, but covers many relevant memory structures within the spirit and scope of the technology as described herein and as understood by one of ordinary skill in the art.

FIG. 1B depicts an example of memory structure 126. In one embodiment, an array of memory cells is divided into multiple planes. In the example of FIG. 1B, memory structure 126 is divided into two planes: plane 141 and plane 142. In other embodiments, more or less than two planes can be used. In some embodiments, each plane is divided into a number of memory erase blocks (e.g., blocks 0-1023, or another amount). In certain memory technologies (e.g. 2D/3D NAND and other types of flash memory), a memory erase block is the smallest unit of memory cells for an erase operation. That is, each erase block contains the minimum number of memory cells that are erased together in a single erase operation. Other units of erase can also be used. In other memory technologies (e.g. MRAM, PCM, etc.) used in other embodiments implementing the solution claimed herein, memory cells may be overwritten without an erase operation and so erase blocks may not exist.

Each memory erase block includes many memory cells. The design, size, and organization of a memory erase block depends on the architecture and design for the memory structure 126. As used herein, a memory erase block is a contiguous set of memory cells that share word lines and bit lines; for example, erase block i of FIG. 1B includes memory cells that share word lines WL0_i, WL1_i, WL2_i and WL3_i and share bit lines BL0-BL69,623.

In one embodiment, a memory erase block (see block i) contains a set of NAND strings which are accessed via bit lines (e.g., bit lines BL0-BL69,623) and word lines (WL0, WL1, WL2, WL3). FIG. 1B shows four memory cells connected in series to form a NAND string. Although four cells are depicted to be included in each NAND string, more or less than four can be used (e.g., 16, 32, 64, 128, 256 or another number or memory cells can be on a NAND string). One terminal of the NAND string is connected to a corresponding bit line via a drain select gate, and another terminal is connected to the source line via a source select gate. Although FIG. 1B shows 69,624 bit lines, a different number of bit lines can also be used.

Each memory erase block and/or each memory storage unit is typically divided into a number of pages. In one embodiment, a page is a unit of programming/writing and a unit of reading. Other units of programming can also be used. One or more pages of data are typically stored in one row of memory cells. For example, one or more pages of data may be stored in memory cells connected to a common word line. A page includes user data and overhead data (also called system data). Overhead data typically includes header information and Error Correction Codes (ECC) that have been calculated from the user data of the sector. The controller (or other component) calculates the ECC when data is being written into the array, and also checks it when data is being read from the array. In one embodiment, a page includes data stored in all memory cells connected to a common word line.

In the example discussed above, the unit of erase is a memory erase block and the unit of programming and reading is a page. Other units of operation can also be used. Data can be stored/written/programmed, read or erased a byte at a time, 1K bytes, 512K bytes, etc. No particular unit of operation is required for the claimed solutions described herein. In some examples, the system programs, erases, and reads at the same unit of operation. In other embodiments, the system programs, erases, and reads at different units of operation. In some examples, the system programs/writes and erases, while in other examples the system only needs to program/write, without the need to erase, because the system can program/write zeros and ones (or other data values) and can thus overwrite previously stored information.

As used herein, a memory storage unit is the set of memory cells representing the smallest storage unit of operation for the memory technology to store/write/program data into the memory structure 126. For example, in one embodiment, the memory storage unit is a page sized to hold 4 KB of data. In certain embodiments, a complete memory storage unit is sized to match the number of physical memory cells across a row of the memory structure 126. In one embodiment, an incomplete memory storage unit has fewer physical memory cells than a complete memory storage unit.

FIG. 2 is a block diagram of example memory system 100, depicting more details of one embodiment of controller 122. As used herein, a flash memory controller is a device that manages data stored on flash memory and communicates with a host, such as a computer or electronic device. A flash memory controller can have various functionality in addition to the specific functionality described herein. For example, the flash memory controller can format the flash memory to ensure the memory is operating properly, map out bad flash memory cells, and allocate spare memory cells to be substituted for future failed cells. Some part of the spare cells can be used to hold firmware to operate the flash memory controller and implement other features. In operation, when a host needs to read data from or write data to the flash memory, it will communicate with the flash memory controller. If the host provides a logical address to which data is to be read/written, the flash memory controller can convert the logical address received from the host to a physical address in the flash memory. (Alternatively, the host can provide the physical address). The flash memory controller can also perform various memory management functions, such as, but not limited to, wear leveling (distributing writes to avoid wearing out specific blocks of memory that would otherwise be repeatedly written to) and garbage collection (after a block is full, moving only the valid pages of data to a new block, so the full block can be erased and reused).

The interface between controller 122 and non-volatile memory die 108 may be any suitable flash interface, such as Toggle Mode 200, 400, or 800. In one embodiment, memory system 100 may be a card-based system, such as a secure digital (SD) or a micro secure digital (micro-SD) card. In an alternate embodiment, memory system 100 may be part of an embedded memory system. For example, the flash memory may be embedded within the host. In other example, memory system 100 can be in the form of a solid-state drive (SSD).

In some embodiments, non-volatile memory system 100 includes a single channel between controller 122 and non-volatile memory die 108, the subject matter described herein is not limited to having a single memory channel. For example, in some memory system architectures, 2, 4, 8 or more channels may exist between the controller and the memory die, depending on controller capabilities. In any of the embodiments described herein, more than a single channel may exist between the controller and the memory die, even if a single channel is shown in the drawings.

As depicted in FIG. 2, controller 122 includes a front-end module 208 that interfaces with a host, a back-end module 210 that interfaces with the one or more non-volatile memory die 108, and various other modules that perform functions which will now be described in detail.

The components of controller 122 depicted in FIG. 2 may take the form of a packaged functional hardware unit (e.g., an electrical circuit) designed for use with other components, a portion of a program code (e.g., software or firmware) executable by a (micro) processor or processing circuitry that usually performs a particular function of related functions, or a self-contained hardware or software component that interfaces with a larger system, for example. For example, each module may include an application specific integrated circuit (ASIC), a Field Programmable Gate Array (FPGA), a circuit, a digital logic circuit, an analog circuit, a combination of discrete circuits, gates, or any other type of hardware or combination thereof. Alternatively, or in addition, each module may include software stored in a processor readable device (e.g., memory) to program a processor for controller 122 to perform the functions described herein. The architecture depicted in FIG. 2 is one example implementation that may (or may not) use the components of controller 122 depicted in FIG. 1A (i.e. RAM, ROM, processor, interface).

Referring again to modules of the controller 122, a buffer manager/bus control 214 manages buffers in random access memory (RAM) 216 and controls the internal bus arbitration of controller 122. A read only memory (ROM) 218 stores system boot code. Although illustrated in FIG. 2 as located separately from the controller 122, in other embodiments one or both of the RAM 216 and ROM 218 may be located within the controller. In yet other embodiments, portions of RAM and ROM may be located both within the controller 122 and outside the controller. Further, in some implementations, the controller 122, RAM 216, and ROM 218 may be located on separate semiconductor die.

Front end module 208 includes a host interface 220 and a physical layer interface (PHY) 222 that provide the electrical interface with the host or next level storage controller. The choice of the type of host interface 220 can depend on the type of memory being used. Examples of host interfaces 220 include, but are not limited to, SATA, SATA Express, SAS, Fibre Channel, USB, PCIe, and NVMe. The host interface 220 typically facilitates transfer for data, control signals, and timing signals.

Back end module 210 includes an error correction code (ECC) engine 224 that encodes the data bytes received from the host, and decodes and error corrects the data bytes read from the non-volatile memory. A command sequencer 226 generates command sequences, such as program and erase command sequences, to be transmitted to non-volatile memory die 108. A RAID (Redundant Array of Independent Dies) module 228 manages generation of RAID parity and recovery of failed data. The RAID parity may be used as an additional level of integrity protection for the data being written into the non-volatile memory system 100. In some cases, the RAID module 228 may be a part of the ECC engine 224. Note that the RAID parity may be added as an extra die or dies as implied by the common name, but it may also be added within the existing die, e.g. as an extra plane, or extra block, or extra WLs within a block. A memory interface 230 provides the command sequences to non-volatile memory die 108 and receives status information from non-volatile memory die 108. In one embodiment, memory interface 230 may be a double data rate (DDR) interface, such as a Toggle Mode 200, 400, or 800 interface. A flash control layer 232 controls the overall operation of back end module 210.

One embodiment includes a writing/reading manager 236, which can be used to manage (in conjunction with the circuits on the memory die) the writing and reading of memory cells. In some embodiments, writing/reading manager 236 performs the processes depicted in the flowcharts described below.

Additional components of system 100 illustrated in FIG. 2 include media management layer 238, which performs wear leveling of memory cells of non-volatile memory die 108. System 100 also includes other discrete components 240, such as external electrical interfaces, external RAM, resistors, capacitors, or other components that may interface with controller 122. In alternative embodiments, one or more of the physical layer interface 222, RAID module 228, media management layer 238 and buffer management/bus controller 214 are optional components that are not necessary in the controller 122.

The Flash Translation Layer (FTL) or Media Management Layer (MML) 238 may be integrated as part of the flash management that may handle flash errors and interfacing with the host. In particular, MML may be a module in flash management and may be responsible for the internals of NAND management. In particular, the MML 238 may include an algorithm in the memory device firmware which translates writes from the host into writes to the memory structure 126 of die 108. The MML 238 may be needed because: 1) the memory may have limited endurance; 2) the memory structure 126 may only be written in multiples of pages; and/or 3) the memory structure 126 may not be written unless it is erased as a block. The MML 238 understands these potential limitations of the memory structure 126 which may not be visible to the host. Accordingly, the MML 238 attempts to translate the writes from host into writes into the memory structure 126. As described below, erratic bits may be identified and recorded using the MML 238. This recording of erratic bits can be used for evaluating the health of blocks and/or word lines (the memory cells on the word lines).

Controller 122 may interface with one or more memory dies 108. In one embodiment, controller 122 and multiple memory dies (together comprising non-volatile storage system 100) implement a solid-state drive (SSD), which can emulate, replace or be used instead of a hard disk drive inside a host, as a NAS device, in a laptop, in a tablet, in a server, etc. Additionally, the SSD need not be made to work as a hard drive.

Some embodiments of a non-volatile storage system will include one memory die 108 connected to one controller 122. However, other embodiments may include multiple memory die 108 in communication with one or more controllers 122. In one example, the multiple memory die can be grouped into a set of memory packages. Each memory package includes one or more memory die in communication with controller 122. In one embodiment, a memory package includes a printed circuit board (or similar structure) with one or more memory die mounted thereon. In some embodiments, a memory package can include molding material to encase the memory dies of the memory package. In some embodiments, controller 122 is physically separate from any of the memory packages.

FIG. 3 is a perspective view of a portion of one example embodiment of a monolithic three-dimensional memory structure 126, which includes a plurality memory cells. For example, FIG. 3 shows a portion of one block of memory. The structure depicted includes a set of bit lines BL positioned above a stack of alternating dielectric layers and conductive layers. For example, purposes, one of the dielectric layers is marked as D and one of the conductive layers (also called word line layers) is marked as W. The number of alternating dielectric layers and conductive layers can vary based on specific implementation requirements. One set of embodiments includes between 108-216 alternating dielectric layers and conductive layers, for example, 96 data word line layers, 8 select layers, 4 dummy word line layers and 108 dielectric layers. More or less than 108-216 layers can also be used. As will be explained below, the alternating dielectric layers and conductive layers are divided into four “fingers” by local interconnects LI (isolation areas). FIG. 3 only shows two fingers and two local interconnects LI. Below and the alternating dielectric layers and word line layers is a source line layer SL. Memory holes are formed in the stack of alternating dielectric layers and conductive layers. For example, one of the memory holes is marked as MH. Note that in FIG. 3, the dielectric layers are depicted as see-through so that the reader can see the memory holes positioned in the stack of alternating dielectric layers and conductive layers. In one embodiment, NAND strings are formed by filling the memory hole with materials including a charge-trapping layer to create a vertical column of memory cells. Each memory cell can store one or more bits of data. More details of the three-dimensional monolithic memory structure 126 is provided with respect to FIG. 4.

FIG. 4 depicts an example 3D NAND structure and shows physical word lines WLL0-WLL47 running across the entire block. The structure of FIG. 4 can correspond to a portion of one of the blocks of FIG. 1B, including bit lines 311, 312, 313, 314, . . . , 319. Within the block, each bit line is connected to four NAND strings. Drain side selection lines SGD0, SGD1, SGD2 and SGD3 are used to determine which of the four NAND strings connect to the associated bit line. The block can also be thought of as being divided into four sub-blocks SB0, SB1, SB2 and SB3. Sub-block SB0 corresponds to those vertical NAND strings controlled by SGD0 and SGS0, sub-block SB1 corresponds to those vertical NAND strings controlled by SGD1 and SGS1, sub-block SB2 corresponds to those vertical NAND strings controlled by SGD2 and SGS2, and sub-block SB3 corresponds to those vertical NAND strings controlled by SGD3 and SGS3.

FIG. 5 illustrates another memory structure that can be used for the structure 126 of FIG. 1A. FIG. 5 illustrates a three-dimensional vertical cross-point structure, the word lines still run horizontally, with the bit lines oriented to run in a vertical direction.

FIG. 5 depicts one embodiment of a portion of a monolithic three-dimensional memory array structure 126 that includes a first memory level 412 positioned below a second memory level 410. As depicted, the local bit lines LBL₁₁-LBL₃₃ are arranged in a first direction (i.e., a vertical direction) and the word lines WL₁₀-WL₂₃ are arranged in a second direction perpendicular to the first direction. This arrangement of vertical bit lines in a monolithic three-dimensional memory array is one embodiment of a vertical bit line memory array. As depicted, disposed between the intersection of each local bit line and each word line is a particular memory cell (e.g., memory cell M₁₁₁ is disposed between local bit line LBL₁₁ and word line WL₁₀). This structure can be used with a number of different memory cell structures. In one example, the particular memory cell may include a floating gate device or a charge trap device (e.g., using a silicon nitride material). In another example, the particular memory cell may include a reversible resistance-switching material, a metal oxide, a phase change memory (PCM) material, or a ReRAM material. The global bit lines GBL₁-GBL₃ are arranged in a third direction that is perpendicular to both the first direction and the second direction. A set of bit line select devices (e.g., Q₁₁-Q₃₁), such as a vertical thin film transistor (VTFT), may be used to select a set of local bit lines (e.g., LBL₁₁-LBL₃₁). As depicted, bit line select devices Q₁₁-Q₃₁ are used to select the local bit lines LBL₁₁-LBL₃₁ and to connect the local bit lines LB₁₁-LBL₃₁ to the global bit lines GBL₁-GBL₃ using row select line SG₁. Similarly, bit line select devices Q₁₂-Q₃₂ are used to selectively connect the local bit lines LBL₁₂-LBL₃₂ to the global bit lines GBL₁-GBL₃ using row select line SG₂ and bit line select devices Q₁₃-Q₃₃ are used to selectively connect the local bit lines LBL₁₃-LBL₃₃ to the global bit lines GBL₁-GBL₃ using row select line SG-3.

Referring to FIG. 5, as only a single bit line select device is used per local bit line, only the voltage of a particular global bit line may be applied to a corresponding local bit line. Therefore, when a first set of local bit lines (e.g., LB₁₁-LBL₃₁) is biased to the global bit lines GBL₁-GBL₃, the other local bit lines (e.g., LBL₁₂-LBL₃₂ and LBL₁₃-LBL₃₃) must either also be driven to the same global bit lines GBL₁-GBL₃ or be floated. In one embodiment, during a memory operation, all local bit lines within the memory array are first biased to an unselected bit line voltage by connecting each of the global bit lines to one or more local bit lines. After the local bit lines are biased to the unselected bit line voltage, then only a first set of local bit lines LB₁₁-LBL₃₁ are biased to one or more selected bit line voltages via the global bit lines GBL₁-GBL₃, while the other local bit lines (e.g., LBL₁₂-LBL₃₂ and LBL₁₃-LBL₃₃) are floated. The one or more selected bit line voltages may correspond with, for example, one or more read voltages during a read operation or one or more programming voltages during a programming operation.

The memory systems discussed above can be erased, programmed/written and read. At the end of a successful programming process, the threshold voltages of the memory cells should be within one or more distributions of threshold voltages for programmed memory cells or within a distribution of threshold voltages (Vts) for erased memory cells, as appropriate. FIG. 6 illustrates example threshold voltage distributions for the memory cell array when each memory cell stores more than one bit of data in a multi-level cell (MLC) format, in this case three bits of data. Other embodiments, however, may use other data capacities per memory cell (e.g., such as one, two, four, or five bits of data per memory cell). FIG. 6 shows eight threshold voltage distributions, corresponding to eight data states. The first threshold voltage distribution (data state) S0 represents memory cells that are erased. The other seven threshold voltage distributions (data states) S1-S17 represent memory cells that are programmed and, therefore, are also called programmed states. Each threshold voltage distribution (data state) corresponds to predetermined values for the set of data bits. The specific relationship between the data programmed into the memory cell and the threshold voltage levels of the cell depends upon the data encoding scheme adopted for the cells. In one embodiment, data values are assigned to the threshold voltage ranges using a Gray code assignment so that if the threshold voltage of a memory erroneously shifts to its neighboring physical state, only one bit will be affected.

FIG. 6 also shows seven read reference voltages, Vr1, Vr2, Vr3, Vr4, Vr5, Vr6, and Vr7, for reading data from memory cells. By testing (e.g., performing sense operations) whether the threshold voltage of a given memory cell is above or below the seven read reference voltages, the system can determine what data state (i.e., S0, S1, S2, S3, . . . ) a memory cell is in.

FIG. 6 also shows seven verify reference voltages, Vv1, Vv2, Vv3, Vv4, Vv5, Vv6, and Vv7. When programming memory cells to data state S1, the system will test whether those memory cells have a threshold voltage greater than or equal to Vv1. When programming memory cells to data state S2, the system will test whether the memory cells have threshold voltages greater than or equal to Vv2. When programming memory cells to data state S3, the system will determine whether memory cells have their threshold voltage greater than or equal to Vv3. When programming memory cells to data state S4, the system will test whether those memory cells have a threshold voltage greater than or equal to Vv4. When programming memory cells to data state S5, the system will test whether those memory cells have a threshold voltage greater than or equal to Vv5. When programming memory cells to data state S6, the system will test whether those memory cells have a threshold voltage greater than or equal to Vv6. When programming memory cells to data state S7, the system will test whether those memory cells have a threshold voltage greater than or equal to Vv7.

In one embodiment, known as full sequence programming, memory cells can be programmed from the erased data state S0 directly to any of the programmed data states S1-S7. For example, a population of memory cells to be programmed may first be erased so that all memory cells in the population are in erased data state S0. Then, a programming process is used to program memory cells directly into data states S1, S2, S3, S4, S5, S6, and/or S7. For example, while some memory cells are being programmed from data state S0 to data state S1, other memory cells are being programmed from data state S0 to data state S2 and/or from data state S0 to data state S3, and so on. The arrows of FIG. 6 represent the full sequence programming. The technology described herein can also be used with other types of programming in addition to full sequence programming (including, but not limited to, multiple stage/phase programming). In some embodiments, data states S1-S7 can overlap, with controller 122 relying on ECC to identify the correct data being stored.

FIG. 7A is a flowchart describing one embodiment of a process for programming that is performed by controller 122. In some embodiments, rather than have a dedicated controller, the host can perform the functions of the controller. In step 702, controller 122 sends instructions to one or more memory die 108 to program data. In step 704, controller 122 sends one or more addresses to one or more memory die 108. The one or more logical addresses indicate where to program the data. In step 706, controller 122 sends the data to be programmed to the one or more memory die 108. In step 708, controller 122 receives a result of the programming from the one or more memory die 108. Example results include that the data was programmed successfully, an indication that the programming operation failed, and indication that the data was programmed but at a different location, or other result. In step 710, in response to the result received in step 708, controller 122 updates the system information that it maintains. In one embodiment, the system maintains tables of data that indicate status information for each block. This information may include a mapping of logical addresses to physical addresses, which blocks/word lines are open/closed (or partially opened/closed), which blocks/word lines are bad, etc.

In some embodiments, before step 702, controller 122 would receive host data and an instruction to program from the host, and the controller would run the ECC engine 224 to create code words from the host data, as known in the art and described in more detail below. These code words are the data transmitted in step 706. Controller 122 (e.g., writing/reading manager 236) can also scramble the data prior to programming the data in the memory.

FIG. 7B is a flowchart describing one embodiment of a process for programming. The process of FIG. 7B is performed by the memory die in response to the steps of FIG. 7A (i.e., in response to the instructions, data and addresses from controller 122). In one example embodiment, the process of FIG. 7B is performed on memory die 108 using the one or more control circuits discussed above (see FIG. 1), at the direction of state machine 112. The process of FIG. 7B can also be used to implement the full sequence programming discussed above. The process of FIG. 7B can also be used to implement each phase of a multi-phase programming process.

Typically, the program voltage applied to the control gates (via a selected word line) during a program operation is applied as a series of program pulses. Between programming pulses are a set of verify pulses to perform verification. In many implementations, the magnitude of the program pulses is increased with each successive pulse by a predetermined step size. In step 770 of FIG. 7B, the programming voltage (Vpgm) is initialized to the starting magnitude (e.g., ˜12-16V or another suitable level) and a program counter PC maintained by state machine 112 is initialized at 1. In step 772, a program pulse of the program signal Vpgm is applied to the selected word line (the word line selected for programming). In one embodiment, the group of memory cells being programmed concurrently are all connected to the same word line (the selected word line). The unselected word lines receive one or more boosting voltages (e.g., ˜7-11 volts) to perform boosting schemes known in the art. If a memory cell should be programmed, then the corresponding bit line is grounded. On the other hand, if the memory cell should remain at its current threshold voltage, then the corresponding bit line is connected to Vdd to inhibit programming. In step 772, the program pulse is concurrently applied to all memory cells connected to the selected word line so that all of the memory cells connected to the selected word line are programmed concurrently. That is, they are programmed at the same time or during overlapping times (both of which are considered concurrent). In this manner all of the memory cells connected to the selected word line will concurrently have their threshold voltage change, unless they have been locked out from programming.

In step 774, the appropriate memory cells are verified using the appropriate set of verify reference voltages to perform one or more verify operations. In one embodiment, the verification process is performed by applying the testing whether the threshold voltages of the memory cells selected for programming have reached the appropriate verify reference voltage.

In step 776, it is determined whether all the memory cells have reached their target threshold voltages (pass). If so, the programming process is complete and successful because all selected memory cells were programmed and verified to their target states. A status of “PASS” is reported in step 778. If, in 776, it is determined that not all of the memory cells have reached their target threshold voltages (fail), then the programming process continues to step 780.

In step 780, the system counts the number of memory cells that have not yet reached their respective target threshold voltage distribution. That is, the system counts the number of memory cells that have, so far, failed the verify process. This counting can be done by the state machine, the controller, or other logic. In one implementation, each of the sense blocks will store the status (pass/fail) of their respective cells. In one embodiment, there is one total count, which reflects the total number of memory cells currently being programmed that have failed the last verify step. In another embodiment, separate counts are kept for each data state.

In step 782, it is determined whether the count from step 780 is less than or equal to a predetermined limit. In one embodiment, the predetermined limit is the number of bits that can be corrected by error correction codes (ECC) during a read process for the page of memory cells. If the number of failed memory cells is less than or equal to the predetermined limit, than the programming process can stop and a status of “PASS” is reported in step 778. In this situation, enough memory cells programmed correctly such that the few remaining memory cells that have not been completely programmed can be corrected using ECC during the read process. In some embodiments, step 780 will count the number of failed cells for each sector, each target data state or other unit, and those counts will individually or collectively be compared to a threshold in step 782.

In another embodiment, the predetermined limit can be less than the number of bits that can be corrected by ECC during a read process to allow for future errors. When programming less than all of the memory cells for a page, or comparing a count for only one data state (or less than all states), than the predetermined limit can be a portion (pro-rata or not pro-rata) of the number of bits that can be corrected by ECC during a read process for the page of memory cells. In some embodiments, the limit is not predetermined. Instead, it changes based on the number of errors already counted for the page, the number of program-erase cycles performed or other criteria.

If number of failed memory cells is not less than the predetermined limit, than the programming process continues at step 784 and the program counter PC is checked against the program limit value (PL). Examples of program limit values include 12, 20 and 30; however, other values can be used. If the program counter PC is not less than the program limit value PL, then the program process is considered to have failed and a status of FAIL is reported in step 788. This is one example of a program fault. If the program counter PC is less than the program limit value PL, then the process continues at step 786 during which time the Program Counter PC is incremented by 1 and the program voltage Vpgm is stepped up to the next magnitude. For example, the next pulse will have a magnitude greater than the previous pulse by a step size (e.g., a step size of 0.1-0.5 volts). After step 786, the process loops back to step 772 and another program pulse is applied to the selected word line so that another iteration (steps 772-786) of the programming process of FIG. 7B is performed.

In general, during verify operations and read operations, the selected word line is connected to a voltage (one example of a reference signal), a level of which is specified for each read operation (e.g., see read reference voltages Vr1, Vr2, Vr3, Vr4, Vr5, Vr6, and Vr7, of FIG. 6) or verify operation (e.g. see verify reference voltages Vv1, Vv2, Vv3, Vv4, Vv5, Vv6, and Vv7 of FIG. 6) in order to determine whether a threshold voltage of the concerned memory cell has reached such level. After applying the word line voltage, the conduction current of the memory cell is measured to determine whether the memory cell turned on (conducted current) in response to the voltage applied to the word line. If the conduction current is measured to be greater than a certain value, then it is assumed that the memory cell turned on and the voltage applied to the word line is greater than the threshold voltage of the memory cell. If the conduction current is not measured to be greater than the certain value, then it is assumed that the memory cell did not turn on and the voltage applied to the word line is not greater than the threshold voltage of the memory cell. During a read or verify process, the unselected memory cells are provided with one or more read pass voltages at their control gates so that these memory cells will operate as pass gates (e.g., conducting current regardless of whether they are programmed or erased).

There are many ways to measure the conduction current of a memory cell during a read or verify operation. In one example, the conduction current of a memory cell is measured by the rate it discharges or charges a dedicated capacitor in the sense amplifier. In another example, the conduction current of the selected memory cell allows (or fails to allow) the NAND string that includes the memory cell to discharge a corresponding bit line. The voltage on the bit line is measured after a period of time to see whether it has been discharged or not. Note that the technology described herein can be used with different methods known in the art for verifying/reading. Other read and verify techniques known in the art can also be used.

In some embodiments, controller 122 receives a request from the host (or a client, user, etc.) to program host data (data received from the host) into the memory system. In some embodiments, controller 122 arranges the host data to be programmed into units of data. For example, controller 122 can arrange the host data into pages, partial pages (a subset of a page), word line units, blocks, jumbo blocks, or other units.

Step 772 of FIG. 7B includes applying a program voltage pulse on the selected word line. Step 774 of FIG. 7B includes verification, which in some embodiments comprises applying the verify reference voltages on the selected word line. As steps 772 and 774 are part of an iterative loop, the program voltage is applied as a series of voltage pulses that step up in magnitude. Between voltage pulses, verify reference voltages are applied. This is depicted in FIG. 7C, which shows program voltage pulses 792, 794 and 796, applied during three successive iterations of step 772. Between program voltage pulses 792, 794 and 796, the system tests the memory cells to determine whether threshold voltages of the memory cells are greater than the respective verify reference voltages by applying the verify references voltages as verify pulses.

FIG. 8 is a flowchart describing a sensing operation performed in order to read data from the memory cells. In step 800, a pass voltage is applied to unselected word lines so that unselected memory cells on a NAND string are conducting, which enables the system to test whether the selected memory cell conducts in response to the read reference voltage. This pass voltage is often referred to as Vread. In step 802, the appropriate read reference voltage, also referred to as Vcgr, is applied to the selected word line. In one example of a system that stores one bit per memory cell, Vcgr=0 v, or a small voltage near 0 v. In step 804, all of the bit lines are pre-charged. In one example embodiment, the bit lines are pre-charged by charging a capacitor in the sense amplifier and then putting the bit line in communication with the charged capacitor so that the bit line charges up. In step 806, the bit line is allowed to discharge, for example, by discharging the capacitor. After a predetermined time period, referred to as the “integration time” or “strobe time” the voltage of the capacitor is sampled to see whether the respective memory cell(s) conducted in step 810. If the memory cell conducts in response to Vcgr, then the threshold voltage of the memory cell is less than Vcgr. If Vcgr=0 v and the memory cell turns on, then the memory cell is in the erased state and the data stored is 1. If Vcgr=0V and the memory cell does not turn on, then the memory cell is in the programmed state and the data stored is 0.

The storage density of a memory circuit such as in FIG. 1B, 3, 4, or 5 can be increased by storing more data states in each of the cells. For instance, FIG. 6 shows a 3-bit per cell example, where each memory cell can store one of 8 different data states. Storing 8 or even more states per cell presents a number of difficulties, as either the different state distributions need to be stored closer together, a larger range of threshold voltages (or “Vt window”) needs to be used, or both. However, programming memory cell states more closely together becomes increasingly complicated, as obtaining sufficiently tight, well-separated distributions can significantly lower performance; and accurate data retention is harder as a smaller amount of threshold voltage drift can make reading the data difficult or even impossible. With respect to increasing the Vt window, going to higher threshold voltages allows for more states to be added at the high Vt end, but at the cost of increased operating voltages, increasing power consumption and possibly shortening device life. Alternately, the Vt window can be increased by extending it further into negative threshold voltages.

In the example of FIG. 6, only the distribution of the lowest, or erased, data state of S0 has a threshold voltage below 0V. Storing more states with negative threshold values can increase the Vt window. This is illustrated in FIGS. 9A and 9B.

FIG. 9A is similar to FIG. 6, but only shows the lowest threshold state S0 and the highest of threshold state SN. The effective Vt window in this example is from around −1V or a few tenths of a volt less on the low side to several volts on the high side (in the 4-6V range, for example, such as 5V), with the other state distributions falling in between these two values. If S0 is the only state whose Vt is below 0V, this can be read by setting the control gate of the memory cell to ground at step 802 in the flow of FIG. 8. FIG. 9B illustrates lowering the bottom end of the Vt window deeper into negative Vt values. In this example, the S0 distribution is at or below Vt=−2.5V to −1.5V (e.g., around −2V or so), adding around 1V to the Vt window and providing additional room for more data states, such as illustrated by S1. However, distinguishing between different data states with negative thresholds has historically been difficult. Therefore, FIGS. 10A-10C present some techniques for sensing negative Vt values.

FIGS. 10A-10C are simplified representations showing a NAND string having only one memory cell connected in series between a source side select gate SGS and a drain side select gate SGD. To simplify the figures, other, non-selected memory cells of the NAND string are not shown but would be biased at a read pass voltage allowing them to conduct for any stored data states. The NAND string is connected on the source end to a source line SRC and on the drain side to a bit line, which is in turn connected to a sense amplifier.

FIG. 10A illustrates a sensing operation for the memory cell using a negative word line voltage CGRV to sense a negative threshold. Aside from the negative word line voltage, the NAND string is biased as is common for reading positive threshold voltage values. The drain and source select gates are set to an on state by applying a sufficiently high voltage along the control lines to their gates, SGD=H and SGD=H, and the source line voltage VSRC is set to ground, VSRC=0V, or other low voltage. The bit line is pre-charged to positive voltage higher than VSRC; for example, VBL can be in the 0.2-1.0V range, such as 0.5V or somewhat lower. The bit line is then discharged by the current Icell, where the rate of discharge is based on the threshold voltage of the memory cell and the word line voltage CGRV on its control gate. After an integration time, the sense amp connected to bit line latches the result. Although this approach can be used for negative Vt states, it requires the use of negative voltages, such as the CGRV=−1.5V illustrated in FIG. 10A. However, negative voltages are typically not used on a memory device as they require additional circuitry to generate and are often difficult to maintain. Additionally, negative voltage levels near or below the −1.5V range are difficult to produce, limiting how deeply the Vt window can be extended downward.

Another approach to sensing negative Vt states, but without negative voltages, is illustrated in FIG. 10B. In FIG. 10B, the biasing of the NAND string is changed to allow a non-negative voltage, such as CGRV=0V, to be used to sense negative Vt states. The select gates (and any non-selected memory cells) of the NAND string are again biased to be on, but now the source line is raised above ground; for example, VSRC can be in the 0.8-1.5V range, such as about 1V or a little higher. This places the source of the selected memory cell at VSRC and allows for negative Vt sensing with a non-negative word line voltage. To discharge the bit line by Icell through the selected memory cell, the bit line is then pre-charged to a level above the source line; for example, VBL can be in the 1.2-1.8V range, such as around 1.5V, placing it a few tenths of a volt above VSRC. After discharging the bit line for a sensing interval, the result is then latched by the corresponding sense amp for the read result. Although this technique allows for negative Vt sensing without negative voltages, it cannot go much deeper into negative Vt values than about −1.1V.

FIG. 10C illustrates another approach that can extend sensing into deeper negative Vt values while only using non-negative voltages. In the sensing arrangement of FIG. 10C, the source line is raised to a voltage level above the bit line voltage level, VSRC>VBL>0V. Rather than determining whether a selected memory cell is conducting by discharging the bit line through the memory cell into the source line, the source line is now discharged through the memory cell into the bit line and sense amp. FIG. 10C illustrates this by the current Icell running upwards towards the bit line, rather than downwards toward the source line as in FIGS. 10A and 10B. For example, the source line can be set in the range VSRC=2.0-2.5V, such as low as 2V, and the bit line voltage, VBL, set a few tenths (e.g., 0.2-0.4V) of a volt less than VSRC. With CGRV=0V, this allows for sensing of a Vt down to about, for example, −1.8V or even further depending on the VSRC and VBL levels. The approach of FIG. 10C is utilized in the following sense amplifier embodiments that can be used for deep negative threshold voltage sensing and techniques for reducing the noise that can occur in such sensing operations.

FIG. 11 illustrates an embodiment of a sense amplifier 1110 that can be used to perform a sensing operation using the approach of FIG. 10C. The sense amplifier 1110 can correspond to one of the sense blocks SB1, SB2, . . . , SBp 150 in FIG. 1A. A selected memory cell 1101 is connected between a source line SRC 1103 and a bit line BL 1105. Other memory cells (e.g. of same NAND string) are also connected between source line SRC 1103 and bit line BL 1105; however, those other memory cells are not depicted in FIG. 11. The sense amplifier 1110 is typically selectively connectable to multiple bit lines through a column decoding circuit not shown in FIG. 11. The selected bit line BL 1105 can be discharged on the path labelled “BL path” through the series connected switches BLC2 1111 and BLC 1113 to the discharge transistor DT 1115, and then on through the discharge transistor DT 1115 to the discharge node SRCGND. The control gate of the discharge transistor DT 1115 is connected in a diode type arrangement to the internal (to the sense amp) bit line node BLI between BLC2 1111 and BLC 1113. When both of BLC2 1111 and BLC 1113 are on, the current flowing to the bit line BL 1105 from the memory cell 1101 can discharge along the BL path to the discharge node SRCGND; and when both of BLC2 1111 and BLC 1113 are turned off, the discharge BL path is cut off and the control gate of DT 1115 is left to float at the level on the node BLI between BLC2 1111 and BLC 1113.

To the right in FIG. 11, a second discharge path labelled “SEN path” allows for the sensing node SEN to also discharge through the switch XXL 1121 to the discharge transistor DT 1115. When XXL 1121 is on, any charge stored on the capacitor Csen 1123 will discharge at a rate determined by the control gate voltage on discharge transistor DT 1115. After discharging over a sensing period, a sensing result based on the level on the node SEN can then be set in the latch 1125 and the shifted out over the data bus DBUS. The SEN node can be pre-charge by the latch 1125.

The voltage levels and timing for the switches in FIG. 11 are controlled by the elements on the memory array such as the read/write circuits 128 and sense blocks SB1, SB2, . . . , SBp 150 in FIG. 1A, here represented by the bias circuitry of control block 1131. A sensing operation, such as a read or verify, is done in two stages. After the initial biasing of the source line SRC, bit line BL, the selected memory and other elements (such as select gates and non-selected memory cells in a NAND embodiment), the switches BLC2 1111 and BLC 1113 are turned on. The bit line is then discharged along the BL path through the discharge transistor. The degree of discharge, or whether there is any current discharged at all, will depend on the word line voltage CGRV on the selected memory cell's control gate and the selected memory cell's threshold voltage Vt. Consequently, the voltage on the node BLI will depend on the memory cell's data state and how this data state corresponds to the read level CGRV biasing the selected memory cell. After the voltage level on the node BLI is sufficiently stabilized, the switches BLC2 1111 and BLC 1113 are turned off, leaving the node BLI, and consequently the control gate of the discharge transistor DT 1115, to float at the level set during the bit line discharge phase.

Once the switches BLC2 1111 and BLC 1113 are turned off and the gate of the discharge transistor DT 1115 is floating at the level set during the bit line discharge phase, the conductivity of the transistor DT 1115 is based on the conductivity of the selected memory cell. In the sense node discharging phase, the switch XXL 1121 is turned on so that the previously charged sense node SEN and the sense node capacitor Csen 1123 can discharge through discharge transistor DT 1115 along the SEN path. After a discharge time, the value at the SEN node can then be captured by latch 1125. As the discharge rate along the SEN path depends on the gate voltage on the discharge transistor DT 1115, which in turn depends on the state of the selected memory cell, the latched value corresponds to the data state. For a memory cell biased as illustrated in FIG. 10C, VCGR=0V is used for sensing the lowest (i.e., most negative) data state, with the VCGR value increased to sense higher Vg states, both less negative Vt states and positive Vt states.

A number of variations in FIG. 11 are possible. For example, rather than having switches BLC2 1111 and BLC 1113 connected in series between BL 1105 and the central SCOM node as shown, one of these can be moved to between the node BLI and the gate of the discharge transistor DT 1115. This arrangement will also allow the level on the control gate of discharge transistor DT 1115 to be set by the voltage level on the BLI node when both switches are on; and close off the BL path and leave the control gate on discharge transistor DT 1115 to float when both are turned off. In another variation, the BL path and the SEN path could discharge through different transistors, but where the gates on the different transistors are tied together. These and other variations can be incorporated in the embodiment of FIG. 11 and other embodiments described below.

To more accurately sense data values, noise during the sensing process should be minimized to the extent practical, particularly when larger number of states are to be stored with the available Vt window. To this end, several techniques can be applied to the sense amp embodiments illustrated in FIG. 11 to provide improved product reliability and performance. Two sources of noise relate to the discharge transistor DT 1115 of FIG. 11, where noise on either the gate of the transistor, or, equivalently node BLI, and noise along the current paths through the transistor can throw off the sensing process.

To reduce noise on the current path through discharge transistor DT 1115, a clamp device and an auxiliary current source, or “keeper current,” can be introduced into the sense amp circuit to clamp the drain voltage of the discharge transistor DT 1115 during sensing. This can help block the possible noise through discharge transistor DT 1115 and provide a current flow through discharge transistor DT 1115 to the node SRCGND. The node SRCGND will typically be a node on a commonly regulated SRCGND line to which the sense amp and other sense amps are connected, so that during a sensing operation all of the connected sense amps may be discharging current into the SRCGND line. The introduction of the auxiliary keeper current helps to remove the critical noise at SRCGND node during sensing.

To reduce noise at the control gate of the discharge transistor DT 1115, a de-coupling capacitor can be introduced to compensate and correct the possible coupling when the switches BLC2 1111 and BLC 1113 switch off to prepare for discharge of sense node. This solution will help to correct the possibly unwanted coupling to the gate of the discharge transistor DT 1115 and provides a more accurate sensing result. The de-coupling capacitor can track operating conditions, such as the temperature, and device corners in order to obtain more accurate sensing results. This can be useful to provide accurate sensing results with temperature dependence and device corners, since the level of how negative a Vt can be sensed may depend on the temperature and device corners.

FIG. 12 includes these immediately above-described elements to reduce noise, as well as other elements that can be incorporated into various embodiments for a sense amp circuit, such as can be incorporated in the sense blocks SB1, SB2, . . . , SBp 150 of FIG. 1A. In FIG. 12, the elements of FIG. 11 are repeated along with a decoupling capacitor Cdecop 1212 connected to the node BLI and a supplemental current source NLO 1218 to help stabilize the SRCGND node during transition to the sensing phase.

More explicitly, FIG. 12 illustrates a memory cell 1201 connected between a source line SRC 1203 and a bit line BL 1205. The memory cell 1201 can be part of a NAND string of charge storing memory cells, such as described with respect to FIGS. 3 and 4, a memory cell based on a phase change memory material (PCM), such as described above with respect to FIG. 5, or of other memory technology. The bit line BL 1205 is connected to the sense amplifier through the decoding circuitry, here represented by the bit line select (BLS) switch 1206. (In this discussion, the switches are generally named according to their control signals from the bias control circuit, such that, for example, the control signal BLS for switch 1206 is also used for its name.)

After the bit line select switch BLS 1206, the bit line BL 1205 is connected to the internal bit line BLI through switch BLC2 1211, and then through switch BLC 1213 to the central comment sensing node SCOM. The node SCOM is connected through the discharge transistor DT 1215 to allow the node SCOM to discharge to SRCGND. Similar to FIG. 11, this provides the discharge path labelled BL path from SRC 1203 through the selected memory cell 1201 to the selected bit line, on through the series connected switches BLC2 1211 and BLC 1213 to the discharge transistor DT 1215, and finally on to SRCGND. The control gate of DT 1215 is again connected to the node at BLI, so that when the switches BLC2 1211 and BLC 1213 are turned off, the control gate of DT 1215 will be left floating at the level on BLI.

To the right on FIG. 12, the SEN node with the capacitor Csen 1223 is connected to the SCOM node through the switch XXL 1221, and then on to discharge transistor DT 1215 to provide the second discharge path (SEN path) from the SEN node to SRCGND for the second of the sensing operations. The SEN node is also connected to the latch 1225 to latch the result of the sensing operation, which is turn connected to the data bus DBUS. Depending on the embodiment, the latch 1225 can include a number of individual latches for use in multi-state reading and writing or for other data operations. The elements of FIG. 12 described so far are largely as described above with respect FIG. 11, except that to simplify FIG. 12 the bias control block (1131 of FIG. 11) to provide the control signals for the various switches is not shown (but is to be included in the device). Some of the waveforms provided to the elements in FIG. 12 are shown in FIG. 13, as described below.

FIG. 12 explicitly shows a number of elements not shown in FIG. 11, but which can be added in various embodiments. A switch NLO2 1207 is connected between SRCGND and a node between BLS 1206 and BLC2 1211, allowing for BL 1205 or BLI to be pre-charged or set to various voltage levels from SRCGND. A switch INV 1216 is connected between the discharge transistor DT 1215, allowing for the sense amp to be selectively isolated from SRCGND, as the SRCGND node may be connected to a line commonly shared by a large number of other sense amps. A switch GRS 1217 is connected in parallel with DT 1215, allowing DT 1215 to be bypassed if, for example, the level on BLI is low so that DT 1215 is off, and the sense amp needs to discharge the DCOM node above DT 1215 to SRCGND. These and various other switches can be added to the sense amp circuit to improve operation and versatility.

The embodiment of FIG. 12 also includes some additional elements that do not directly enter into the main sensing operations described here, but can also add to its versatility. A switch BIAS 1204 can connect the bit line to a level BLBIAS that can be used in biasing a selected bit line for various memory operations. Also, another path to the central SCOM node is provided through a switch BLX 1241 (and possibly additional switches) to a high sense amp voltage VHSA. Although not used in the sensing operations mainly described here (where all data states are sensed by discharging the SRC line 1203 through the memory cell 1201 into the sense amp), more standard sensing operations as illustrated in FIG. 10A (where the sense amp/bit line discharge through the memory cell 1201 into SRC 1203) could use the switch BLX 1241. For example, rather than sense all states as illustrated by FIG. 10C, successively raising CGRV from 0V through the various read valued, the approach of FIG. 10C could be used for negative Vt states and then switch to the approach of FIG. 10A for the non-negative Vt states that do not require a negative CGRV value when using the approach of FIG. 10A.

As described above with respect to FIG. 11, the sense amplifier arrangement of FIG. 12 can be used to perform a sensing operation on a selected memory cell by a first phase using the first discharge path “BL path” to discharge the source line 1203 through the selected memory cell 1201 and on through the discharge transistor DT 1215 to SRCGND. This will set the node at BLI, and the control gate on DT 1215, to a voltage level dependent on the data state of the memory relative to the voltage level CGRV on the corresponding word line. Once the level on BLI node is stabilized, the switches BLC2 1211 and BLC 1213 are turned off, leaving the control gate of DT 1215 to float, and the conductivity of DT 1215 to be determined by the conductivity of the selected memory cell 1201. In the second phase, the switch XXL 1221 is turned on to discharge the pre-charged node SEN through DT 1215 to SRCGND at a rate based on the conductivity of DT 1215, which is in turn based on the conductivity of the memory cell 1201. After a discharge period, the level on SEN is captured by the latch 1225 to give the sensing result.

To reduce noise on the BLI node, and the gate of discharge transistor DT 1215, when the switches BLC2 1211 and BLC 1213 are turned off during the transition, the de-coupling capacitor Cdecop 1212 is introduced. This capacitor helps to compensate and correct for the possibly of unwanted coupling to the gate of the discharge transistor DT 1215 and provide a more accurate sensing result. The lower plate of Cdecop 1212 is connected to the BLI node, with the upper plate connected to a level BLI BST that can allow the de-coupling capacitor Cdecop 1212 to track the operating conditions, such as temperature, and device corners in order to obtain a more accurate sensing result. In some embodiments, Cdecop 1212 can be implemented as a transistor with both its source and drain connected to the BLI node and its control gate connected to the level BLI BST.

Another source of noise during the transition between phases and the subsequent discharging of the SEN node can come from noise in the SRCGND level, where the SRCGND line will typically be shared by a large number of sense amps that will concurrently be dumping current into the SRCGND line. A supplemental current source through the switch NLO 1218 is connected to a sense amp voltage LVSA to provide a keeper current through the discharge transistor 1215. A clamp device DCL 1219 clamps the drain voltage (at node DCOM) of discharge transistor DT 1215 during the sensing. These devices help block the possible noise through discharge transistor DT 1215 and keep a constant current through to the commonly regulated node SRCGND. This can help remove the detrimental noise at SRCGND node during sensing.

FIG. 13 illustrates waveforms for the control signals from the biasing circuitry for some of the control signals for FIG. 12 in one embodiment for a sensing operation. The waveforms are marked at times t0-t10, where: t0-t3 is preparatory period; t3-t6 is the first phase where the source line SRC 1203 is discharged into the sense amp and the level on the control gate of the discharge transistor DT 1215 set; t6-t8 is the transition between phases; t8-t9 is the second phase when the SEN node is discharged along the second discharge, or SEN, path; and t9-t10 is the strobe period when the value on the SEN node is latched.

The control signals for some of the devices in FIG. 12 are not included in the waveforms of FIG. 13. INV 1216 in on, and GRS 1217 is off during the all of the shown period. The bit line select switch BLS 1206 is on for the whole period, or at least until the first phase concludes at t6. As discussed above, the switches BIAS 1204 and BLX 1241 are not active in the sensing operation described with respect to FIG. 13 and would both be off. DCL 1219 acts as a voltage clamp for the node DCOM and has its gate set for this purpose.

Starting at t0 for FIG. 13, BL 1205, the nodes BLI, SCOM, SEN and SRCGND are all low, as are the control signals on NLO2 1207, BLC2 1211, BLC 1213, XXL 1221, the CLK signal to the plate of Csen 1223, and NLO. Between t0 and t1, the array is biased. This can include setting the SRC line 1203, the selected and unselected word lines, select gates, or other levels needed to bias a selected memory cell 1201, depending on the architecture of the array.

Between t1 and t2, the initial levels for the sense amp are set. The SRCGND line is raised to an initial high value and NLO2 1207 is turned on, as is BLC2 1211. This sets the values on BL 1205 and the node at BLI high. Once the bit line and internal bit line are set, between t2 and t3, NLO2 is turned off and SRCGND is lowered to the level used during the following discharge phases.

The first discharge phase along the first discharge path labelled BL path in FIGS. 11 and 12, when the source line SRC 1203 discharges through the selected memory cell 1201 into the sense amp, begins at t3 when BLC 1213 is turned on, connecting the central common sense node SCOM to BLI. The bit line BL 1205 and BLI begin to discharge between t3 and t4, while SCOM charges up. The level on all these three top traces will depend on the conductivity of the selected memory cell, where HC is a highly conductive cell, MC a cell of middle conductivity, and NC a non-conducting cell. As shown, BL, BLI and SCOM will stabilize at t4 with the high conductively cell highest, the non-conducting cell lowest, and the intermediate state in the middle. To prepare for the next phase, at t4 the SEN node is pre-charged, which can be done from the latch 1225, followed by raising the CLK signal to Csen 1223 at t5, which further raises the level on SEN. By t6, the level on BLI (and the control gate of DT 1215) is stabilized at a level based on the conductivity of the selected memory cell and the SEN node is pre-charged, setting the conditions for the second discharge phase.

At t6, BLC2 1211 and BLC 1213 are turned off, isolating the BLI node, so that from t7 on, BLI is floating (represented by the broken lines) at a level based the memory cell's conductivity. This cuts off the discharge path from the source line SRC 1203 and causes the bit line 1205 to go high, where it will stay for the rest of the process, and SCOM to discharge through the discharge transistor DT 1215 and bounce about. This also results in coupling noise on BLI and the gate of DT 1215, as illustrated in FIG. 13 by the jagged outlining of the BLI levels between t6 and t8. The decoupling capacitor Cdecop 1212 is used to help correct for this noise, where the level BLI BST on the upper plate of Cdecop 1212 can track temperature and device corners to provide a more accurate sensing result.

The fluctuations on the SCOM node and the BLI node also introduce noise on SRCGND, which can be very sensitive to noise, as shown by the jagged outline of SRCGNE between t6 and t7. To help remove this noise, the supplemental current from NLO 1218 and the clamp DCL 1219 to keep the level at DCOM help to stabilize the SRCGND node and the SCOM node. As shown on the bottom trace, NLO 1218 is turned on to provide the supplemental keeper current at t7.

At t6, the SEN node has been pre-charged and the control gate of DT 1215 and SRCGND have be stabilized. XXL 1221 is then turned on to discharge the SEN node. The transition in XXL 1221 can again introduce noise for SRCGND, which the keeper current from NLO 1218 will also help stabilize. When XXL 1221 turns on at t8, SCOM and SEN begin to discharge at a rate determine by the gate voltage on DT 1215, which was in turn set by the conductivity of the memory cell. As shown, between t8 and t9 the HC state discharges most rapidly and the NC state shows almost no discharge, while the MC state falls in the middle. At t9-t10, the level on SEN is latched, after which the sensing operation is complete.

FIG. 14 is a flowchart describing one embodiment of a process for a sensing operation for the sense amplifier of FIG. 12 using the waveforms of FIG. 13. Beginning at step 1401, the selected memory cell 1201, source line 1203, and any other array elements (select gates, non-selected memory cell on the same NAND string, etc.) are biased in preparation for the sensing operation. This corresponds the t0-t1 section of FIG. 13. In step 1403, the bit line BL 1205 and internal bit line BLI are charged up, corresponding to the period of t1-t3 of FIG. 13. The first discharge phase then begins in step 1405.

In step 1405, the switch BLC 1213 is turned on and the source SRC 1203 begins the first discharging phase through the selected memory cell 1201 along the first discharge path (BL path), eventually stabilizing at a level depending on the conductivity of the selected memory cell 1201. The level on BLI during this process is also the level on the control gate of the discharge transistor DT 1215, corresponding to step 1407. Steps 1405 and 1407 are during the period t3-t6 of FIG. 13.

The SEN node is pre-charged at step 1409. In the embodiment of FIG. 13, this occurs during the period t4-t6, during the first discharge phase along the BL path. Other embodiments can have this step earlier or later, as long as the SEN node is prepared for subsequent discharge along the SEN path at step 1415.

Steps 1411 and 1413 are part of the transition between the two phases, corresponding to the period t6-t8 in the embodiment of FIG. 13. At step 1411, the BL path for discharge is cut off and the control gate of DT 1215 is set to float at the level on BLI by turning off of the switches BLC2 1211 and BLC 1213. The capacitor Cdecop 1212 helps to reduce the noise on BLI, where having the upper plate connected to the level BLI BST can help with variations due to operating conditions or process corners. At step 1413, the auxiliary keeper current from NLO 1218 begins, helping to stabilize SRCGND.

The second discharge phase for the second discharge path, the SEN path, corresponding to the period t8-t9, begins at step 1415. The switch XXL 1221 is turned on and the SEN node discharges through DT 1215, whose control gate was set based on the conductivity of the selected memory cell 1201 at step 1407. For the embodiment of FIG. 13, the keeper current is kept running during this period to keep SRCGND noise down. At step 1417, the level on SEN is latched to provide the sensing result and concluding the sense operation.

The discussion now considers an additional set of embodiments for a sense amplifier circuit, such as can be used for determining the data states of non-volatile memory cells. These additional embodiments can include many of the components of the embodiments described above, but can also include additional elements. More specifically, these additional embodiments include a first path and a second path between a selected bit line and the SEN node to which the sensing capacitor, C_(SEN), is attached. The first path is used for performing sensing operations in a first mode, in which current conducted through a selected memory cell flows into the sense amplifier and is conducted along the first path to the SEN node to charge up the sensing capacitor. The second path is used for conventional sensing operation in which charge on the sensing capacitor is conducted along the second path and discharged through a selected memory cell. In both modes, the voltage level on the SEN node can then be used to determine the data state of the selected memory cell.

The first current path, used when the sensing is based on currently flowing into the sense amplifier, uses PMOS transistors for conducting current from the bit line to the sensing node. The second current path, used when the sensing is based on currently flowing out of sense amplifier, uses NMOS transistors for conducting current from the sensing node to the bit line. The PMOS devices of the first path are in parallel with the NMOS devices of the second current path. Both sensing directions can be achieved by use of a different set of control signals to bias the sense amplifier, so that the sense amplifier is capable to handle the current of a selected memory cell in both directions.

FIG. 15 is a circuit diagram of an additional embodiment of a sense amplifier operable in a first mode and a second mode. Relative to FIGS. 11 and 12, FIG. 15 repeats a number of sense amplifier elements and includes a number of elements that can be included in the embodiments of FIGS. 11 and 12, but where not shown there in order to simplify the discussion given above. FIG. 15 adds several elements, providing the first current path that can used to charge the SEN node when operating in the first sensing mode in which the SEN node and sensing capacitor are charged from a selected memory cell. A number of the elements (e.g., NLO2 1207, BLC2 1211, BLI BST 1212, DCL 1219, DT 1215, GRS 1217, INV 1216) illustrated in FIG. 12 are not included in FIG. 15 and following figures, but alternate embodiments can also include some or all of these elements into the embodiments described with respect to FIG. 15 and subsequent figures.

More specifically, starting on the left hand side, FIG. 15 illustrates a memory cell 1501 with its source side connected to a source line CELSRC 1503 and with its drain side connected to a bit line that connects to the sense amplifier at the bit line node BL 1505. The node BL 1505 can be biased by a level BLBIAS through a switch BIAS 1504, here implemented as an NMOS device. The bit line node BL 1505 can be connected to an internal bit line BLI by way of a bit line select switch BLS 1506, again implemented as an NMOS device. Depending on the embodiment, internal bit line BLI can be connected to one or multiple bit lines BL 1505 and the bit line select switch BLS 1506 can used to connect the sense amp to a selected memory cell on a selected bit line for sensing operations, for both data read and program verify and to also set the bias level along the bit line to programming bias levels, such as program inhibit and program enable. Depending on the embodiment, the memory cell 1501 can be an individual memory cell or one or a number of memory cells connected between CELSRC 1503 and BL 1505, such as in a NAND string. A single word line 1502 is shown in FIG. 15 and would use to bias the memory cell, such as to a sensing voltage for a read or verify operation or to apply a programming voltage to a selected memory cell, where, in the case of a NAND string for example, WL 1502 is representative of the set of word lines and selected gate lines used to bias NAND string.

BLI is connected to the common node SCOM of the sense amplifier through a bit line clamp switch BLC 1513, again implemented as an NMOS device, and the SCOM node is in turn connected to the sense node SEN though the NMOS device of switch XXL 1521. The sensing capacitor Csen 1523 is connected to the SEN on its top (as represented here) plate and to the control signal CLKSA on its bottom plate. The sensing node SEN is connected to the data latch 1525. Depending on the embodiment, each sense amplifier can be connected to a number of different data latches, such as can be used for different bits stored on a memory cell when used with a multi-level cells (MLC) memory. The common node SCOM is also connectable to the level VHSA though the NMOS device of switch XXL 1541 and to the level SRCGND thought the NMOS device of switch NLO 1518.

Other elements in FIG. 15, not represented in FIGS. 11 and 12, can include the PMOS device of switch 1563, through which BLX 1241 is connectable to VHSA, and NMOS device of switch 1561, though which BLX 1241 is connectable to level SRCGND. As shown in FIG. 15, both of switches 1563 and 1561 have their control gates connected to the control signal INV_S so that when INV_S is high the NMOS of 1561 will be on and the PMOS of 1563 will be off, and when INV_S is low the NMOS of 1561 will be off and the PMOS of 1563 will be on. The devices 1561 and 1563 can be used to set the bit line BL 1505 to a program inhibit or a program enable level during a programming operation, for example.

Between the sensing node SEN and the latch 1525 is an NMOS device switch BLQ 1571, where the internal sensing bus line between BLQ 1571 and latch 1525 is labelled SBUS. The internal sensing bus line SBUS is also connectable to the level VHLB by NMOS device switch LPC 1573 that can be used to pre-charge the line SBUS. SBUS is also connected to the level VLOP through the series connected NMOS devices of switch STB 1567 and 1565, whose gate is connected to the SEN node. The NMOS devices 1565 and STB 1567 can be used to discharge the SBUS mode during a sensing operation. FIG. 15 also includes a control block 1531 that includes the bias circuitry configured to provide the control signals used by the sense amplifier during its operation.

The elements of FIG. 15 described so far are all either used in the second sensing mode, in which the SEN node is discharged through a selected memory cell, or shared between the second mode and the first mode, in which the SEN node is charged through the selected memory cell. As described in more detail below, in the second, traditional sensing mode, the SEN node is discharged through selected memory cell to BL 1505 through the NMOS devices BLC 1513 and XXL 1521, which operate as series connected source follower devices. To provide a first current path between BL 1505 and SEN for use in the first sensing mode, in which the SEN node is charged through the selected memory cell 1501 from CELSRC 1503, several PMOS devices are added in parallel with the NMOS devices of the second current path. These additional PMOS devices are configured to be source follower devices when current flows in the reverse direction from BL 1505 to SEN.

More specifically, in parallel with the NMOS device BLC 1513 between BLI and SCOM is the PMOS device PBLC 1514. Between SCOM and SEN, a PMOS PXXL 1522 is connected in parallel with the NMOS XXL 1521. A PMOS switch PBLX 1542 is also connected in parallel with NMOS BLX 1541 for use in the first sensing mode. Before describing the operation of the circuit of FIG. 15 in the first sensing mode, a brief description of the second sensing mode is given.

In a traditional sensing operation (second sensing mode) of a sense amplifier, a selected memory cell 1501 is biased by setting the word line WL 1502 to a read voltage. The read voltage is applied to the word line 1502 of the selected memory cell and, in the case of a NAND string, would also include biasing the non-selected memory cells and select gates to be conducting. The degree to which the selected memory cell 1501 conducts depends on the level of the read voltage applied to the word line of the selected memory cell relative the threshold voltage of the memory cell, which in turn depends on the data state programmed to the memory cell.

To perform a sensing operation, either a data read or a program verify in the second mode, the SEN node and sensing capacitor Csen 1523 are pre-charged. The SEN node is pre-charged and then discharged along the path from SEN to BL 1505 through XXL 1521 and BLC 1513 at a rate dependent on the conductivity of the memory cell 1501, which in turn depends on the read voltage applied to WL 1502 relative to the threshold voltage of the memory cell 1501. The voltage level on SEN and Csen 1523 determines the level on the gate of 1565, which determines the conductivity of 1565. The voltage on SBUS can also be pre-charged through LPC 1573. After allowing the SEN node to discharge for some interval of time, the strobe transistor STB 1567 is turned on for a sensing, or strobe, interval to allow SBUS to discharge at a rate dependent of the SENS voltage level. Depending on the level on the internal bus line SBUS at the end of the strobing interval, either a 0 or a 1 sensing result is latched into the latch 1525 based on whether the SBUS voltage level is above or below a reference level.

FIG. 16 is a flowchart presenting an embodiment for a second mode sensing operation in which the sensing node in a sense amplifier circuit is discharged through a selected memory cell. At step 1601 the SEN node is pre-charged. For example, SEN can be pre-charged though BLQ 1571 and LPC 1573 from the VHLS level, after which BLQ 1571 and LPC 1573 can be turned off. The SBUS line can then be pre-charged through LPC 1573 from the VHLS level at step 1603. The memory cell (or, in the case of NAND memory, the NAND string) is biased for a sensing operation at step 1605, where, depending on the embodiment, step 1605 can be performed before, after, or currently with steps 1601 and 1603.

Once the sense amplifier is biased to its pre-charge levels at steps 1601 and 1603 and the selected memory cell 1501 (or NAND string) is biased at step 1605, the SEN node is discharged along the second current path through XXL 1521 and BLC 1513 to BL 1505 at step 1607. The rate at which the SEN discharges will depend on the conductivity of memory cell 1501, which, in the case of a NAND memory cell, in turn depends on the level of the read voltage applied to word line 1502 relative to the threshold voltage of the memory cell 1501. For example, if the read voltage is less than the memory cell's threshold voltage memory cell 1501 will not conduct and SEN will not discharge. As the gate of 1565 is connected to the SEN node, the voltage on SEN will determine the conductivity of 1565. After allowing an interval for SEN to discharge, at step 1609 a strobe signal of the control signal to STB 1567 is asserted for a strobe interval allowing SBUS to discharge an amount determined by the voltage level on SEN. At the end of the strobe interval, the level on SBUS relative to a reference level is latched into the latch 1525 as either a 0 or 1 to determine the result of the sensing operation.

FIG. 17 illustrates the discharge of the SEN node through a selected memory cell 1501 to CELSRC of step 1607 of FIG. 16 and the strobe on STB 1567 of step 1609 of FIG. 16. To pre-charge the SEN node at step 1601 in the second sensing mode, the switches XXL 1521, PXXL 1522, and STB 1567 are all turned off and switch BLQ 1571 and LPC 1573 are turned on. This provides a path from VHLB to SEN, allowing the SEN node and the sensing capacitor Csen 1523 to be pre-charged from the VHLB level. (This pre-charge of the SEN node can be performed as illustrated in FIG. 18, where it is also included in the first sensing node.) To pre-charge the SBUS line at step 1603, the switch BLQ 1571 is turned off, the switch STB 1567 is off, and the switch LPC 1573 is on, allowing the SBUS line to charge from the level on VHLB. (The pre-charge of SBUS can be performed as illustrated below in FIG. 20, where it is also included in phase 3 of the first sensing node.)

Once the SEN node and SBUS are pre-charged, the SEN node is discharged through the selected memory cell 1501. The selected memory cell 1501 is biased by setting a read voltage on the word line 1502 and setting the source line CELSRC 1503 to low voltage level. In the case of a NAND string, the non-selected memory cells and selected gates are biased to be conductive. The degree to which the selected memory cell 15011 will conduct will depend on the threshold voltage of the selected memory cell 1501 and the level of the read voltage along word line WL 1502 relative to this threshold value. The series connected, cascaded NMOS switches XXL 1521 and BLC 1513 are on, along with the bit line select switch BLS 1506, with the parallel connected PMOS devices PXXL 1522 and PBLC 1514 of first current path turned off, and the other switches out SCOM (NLO 1518, BLX 1541, PBLX 1542) turned off. This provides the second current path through the source followers of XXL 1521 and BLC 1513 for the SEN node, and the sensing capacitor Csen 1523 to discharge at step 1607 through the selected memory cell 1501 to CELSRC 1503. This is as illustrated by the arrow, where the amount of discharge (or lack of discharge) will depend on the level of the read voltage on WL 1502 relative to the threshold voltage of the selected memory cell 1501.

After a sensing interval for the discharge time, a strobe pulse is applied at STB 1567, corresponding to step 1609. The voltage level on NMOS 1565 is determined by the voltage level on SEN, so that the degree to which SBUS will (or will not) discharge (as represented the arrow to VLOP) will reflect the relationship of the read voltage to the selected memory cells data state as reflected in its threshold voltage. The latching of the level on SBUS at step 1611 can then be performed.

FIGS. 18-21 illustrate an embodiment for a sensing operation for the sense amplifier circuit of FIG. 15 when operating in the first sensing mode, when the sensing node SEN and sensing capacitor Csen 1523 are charged though the selected memory cell 1501 along the first current path from BLI to SEN. FIG. 21 is a flowchart corresponding to the phases illustrated in FIGS. 17-20.

FIG. 18 illustrates a pre-charge phase, or phase 1, for a sensing operation in the first sensing mode. In the pre-charge phase the bit line BL 1505 and source line CELSRC 1503 are pre-charged to a target voltage V(CELSRC). For a NAND based embodiment, this will also raise the channel of the NAND string to V(CELSRC). As shown on the left of FIG. 17, this can be accomplished by setting BLBIAS to V(CELSRC) and turning on the switch BIAS 1504 to set BL 1505 to V(CELSRC) and setting CELSRC 1503 to V(CELSRC). To limit this pre-charging to these elements and the selected memory cell 1501, the switches BLC 1513, NLO 1518, XXL 1521, PXXL 1522, BLX 1541 and PBLX 1542 are all turned off (as indicated by the X through these devices) by their biasing control signals from control block 1531. The INV_S level can also be taken high at this point (or other point prior to phase 2) to turn off PMOS 1563 and turn on 1563.

As shown on the right hand side of FIG. 18, the sensing node SEN and sensing capacitor Csen 1523 are pre-charged from VHLS by turning on LPC 1573 and BLQ 1571. As STB 1567, XXL 1521 and PXXL 1522 are all off, the charge from VHLS accumulates on SEN and sensing capacitor Csen 1523.

FIG. 19 illustrates a settling phase, or phase 2, in which the voltage levels on the sense amplifier circuit are stabilized for a sensing operation in the first sensing mode. As illustrated by the arrow on the left hand side of FIG. 18, the bit line is settled by use of the cascaded PMOS devices PBLC 1514 and PBLX 1542. In more detail, the level on CELSRC 1503 is maintained at the first mode sensing level of ˜2 v, but, relative to FIG. 18, the switch BIAS 1504 is now turned off. As in FIG. 18, BLC 1513, NLO 1518, and the parallel pair of XXL 1521 and PXXL 1522 are all off. Also, as in FIG. 18, BLX 1541 is off and INV_S is high, so that NMOS 1561 is on and PMOS 1563 is off. PBLX 1542 is now on, providing a path from SCOM to SRCGND. As PBLC 1514 is also on, this provides a current path as indicated by the arrow from BL 1505 through the cascaded source follower PMOS devices PBLC 1514 and PBLX 1542 to SRCGND, allowing BL to settle down.

With respect to the SEN node, in the stabilization phase (phase 2) the SEN node is discharged to ˜1V, with the local threshold voltage variation of NMOS 1565 compensated for in the result of the discharge. As in FIG. 18, the SEN node remains cut off from the SCOM node by having both XXL 1521 and PXXL 1522 off. The SEN node having been pre-charged as in FIG. 18, LPC 1573 is now off while BLQ 1571 remains on. To discharge the SEN node, STB 1567 is on. As SEN is connected to the gate of NMOS 1565, and both of BLQ 1571 and STB 1567 are biased to be on, NMOS 1565 is basically a diode connected transistor with the SEN being applied to the gate of NMOS 1565. This will allow SEN to discharge to a level determined by the threshold voltage Vth of NMOS 1565. If the lower plate of the sensing capacitor is at ˜1V, this results in the sense node SEN will discharge to VLOP+Vth=˜1V, cancelling or compensating the local Vt variations between the different sense amplifies on the die.

FIG. 20 illustrates a SEN node charging phase, or phase 3, of an embodiment for the first sensing mode. Relative to phase 2 in FIG. 19, PXXL 1522 is now turned on and BLQ 1571 is turned off. The level of CLKSA is stepped down from ˜1V to 0V. As the source line CELSRC 1503 is at a relative high sensing level (−2 v), current will flow from the source lines and through the selected memory cell 1501 to BL 1505 at a rate dependent on the read voltage applied to the word line WL 1502 relative to the threshold voltage level of the selected memory cell 1501. (As before, in the case of a NAND embodiment, the non-selected memory cells and select gates of the selected NAND string will be biased to be on.) The first current path of the cascaded PMOSs PBLC 1514 and PXXL 1522 (as illustrated by the arrow from the selected memory cell 1501), where the transistors act as source followers and let the SEN node and the upper plate of the sensing capacitor Csen 1523 accumulate charge at a rate based on the conductance of the selected memory cell 1501. Depending on the level of current Icell through the selected memory cell, SEN may stay at ground (0V), rise from ground slowly, or rise rapidly. After an accumulation sensing interval, PXXL 1522 can be turned off to trap the accumulated charge on SEN.

In phase 3, the SBUS line is also pre-charged. With BLQ 1571 and STB 1567 off, LPC 1573 is turned on. This allows for SBUS to be pre-charged form VHLB. Once SBUS is pre-charged, LPC 1573 can be turned off.

FIG. 21 illustrates the strobe phase (phase 4) of the first sensing mode, in which the result is stored in the latch 1525. As illustrated in FIG. 20, charge is accumulated on the SEN node over the sensing interval during phase 3, where the amount of accumulated charge is based on the conductivity of the selected memory cell 1501 in response to the applied read voltage. As the sensing node SEN is connected to the gate of the NMOS 1565, the amount of accumulated charge on SEN will determine the conductivity of NMOS 1565. The process of discharging the SEN node in phase 2 (FIG. 19) cancels the threshold voltage Vth of NMOS 1565 so that the voltage level accumulated on SEN will have the Vth variation removed. By applying a strobe (briefly asserting a control signal for a sensing interval) to turn on STB 1567, SBUS will discharge at a rate based on the voltage level V(SEN) on SEN, which is in turn based on the data state stored in the selected memory cell 1501 relative to the read voltage applied to the word line 1502. If V(SEN)=(Icell*tSense)/C, where C is capacitance of Csen 1523, is greater than a reference value (such as 1V for the process described here), SBUS is discharged. The result of the sensing can then be latched into latch 1525.

Referring back to FIG. 19 and the settling phase, or phase 2, in which the voltage levels on the sense amplifier circuit are stabilized for a sensing operation in the first sensing mode, this phase included settling the bit line BL 1505 by use of a path through SCOM, PBLX 1542 and the switch of NMOS 1561 to SRCGND. The gate of NMOS 1561 is controlled by the signal INV_S. In FIG. 19, INV_S is high so that the NMOS 1561 is on. In a program operation, the value of INV_S can be set based on the result of a preceding program verify operation. When INV_S=high, this indicates that the selected memory cell failed the preceding program verify and that a subsequent verify operation will need to be performed, so that the bit line BL 1505 will be discharged as part of this subsequent verify by setting INV_S high.

When INV_S=low, this indicates that the selected memory cell has passed a preceding program verify and that a subsequent verify operation need not be performed. The memory cell can be locked out from further verify, in which case the bit line BL 1505 need not be discharged for a subsequent verify, saving on power. If INV_S is low, the NMOS 1561 will be off, cutting off the path from bit line BL 1505 through SCOM, PBLX 1542 and the switch of NMOS 1561 to SRCGND. In some modes of operation, such as “no lock-out”, it can be useful to still be able the discharge BL 1505 to SRCGND. To able be able to discharge BL 1505 independently of the INV_S level and without using the PBLX 1542 path, an additional PMOS path between SCOM and SRCGND can be added in parallel with NLO 1518. FIG. 22 illustrates such an alternate embodiment.

FIG. 22 is a circuit diagram of an alternate embodiment of a sense amplifier operable in a first mode and a second mode. Relative to FIG. 15, FIG. 22 adds the PMOS PNLO 1519 in parallel with the NMOS NLO 1518 between SCOM and SRCGND. This provides a path between BLI and SRCGND through the series connected PMOS devices PBLC 1514 and PNLO 1519 operating as source follower devices. This provides a path through which BL 1505 can be discharged to SRCGND regardless of the level on INV_S.

FIG. 22 is also marked to illustrate an alternate embodiment for the process illustrated in FIG. 19 for the settling phase, or phase 2, in which the voltage levels on the sense amplifier circuit are stabilized for a sensing operation in the first sensing mode. On the right had side, the SEN node is again discharged to VLOP+Vth=˜1V as described above with respect to FIG. 19. To settle down BL 1505, PBLC 1514 is now cascaded with PNLO 1519 so that from SCOM the current path is now though PNLO 1519 to SRCGND.

The control signals for biasing the elements of the sense amplifier circuit of FIGS. 15 and 22 are provided by the control block 1531. These control signals can include both the gate voltages for the various transistors and also the values applied at lines such as INV_S and VLOP. FIGS. 23 and 24 illustrate embodiments for some elements of the bias-voltage generation circuitry of the control block 1531.

FIG. 23 is a diagram of an embodiment of a bias-voltage generation circuit that can provide control gate voltages for some of the PMOS elements of the sense amplifier embodiments in FIGS. 15 and 22. More specifically, the circuit of FIG. 23 can provide bias levels for the PMOS devices of PBLC 1514, PXXL 1522, and PBLX 1542 also a CELSRC level for use in the first sensing mode described with respect to FIGS. 18-22. In the first sensing mode of FIGS. 18-22, as the current from a selected NAND string or memory cell flows into the sense amplifier (such as is illustrated with respect to FIG. 10C), the usual roles of the source and drain are reversed and the bit line serves the role normally filled by source line. Because of this, the biasing circuit of FIG. 22 references the bias levels relative to a target bit line level, VBL_TARGET. Other voltages, such as the read voltage applied to the word line of a selected memory cell, can similarly be biased relative to this target bit line level, much the way in which the levels of a traditional sensing operation (in which current flows from the bit lines) are referenced relative to the source line's level.

The target bit line voltage VBL_TARGET for BL 1505 is received at the node 2301. The value of VBL_TARGET will vary depending on the embodiment, but a typical value can be around 2V, such as some tenths of a volt less to perhaps a volt higher. Relative to VBL_TARGET, the target level of CELSRC for FIGS. 18-22 will somewhat higher to reflect a voltage drop across the selected memory cell and the bit line and other elements (such as non-selected memory cells and select gates of a NAND string) between source line CELSRC 1503 and BL 1505. The target voltage for CELSRC is provided from a node 2303, where the voltage drop across the selected memory cell is emulated by the voltage source 2305 and the voltage drop along the bit line is emulated by the voltage source 2307.

The target voltage levels of PBLC, PBLX, and PXXL are all lower than VBL_TARGET and are supplied from nodes between 2301 and the low voltage level on the sense amplifier of VSSSA. (As noted above, switches and their control signals are similarly named, so that PBLC, for example, is used for both the PMOS transistor PBLC 1514 and the control signal applied to its control gate.) A diode connected PMOS 2311 connected the VBL_TARGET node 2301 and, though current source 2323, VSSSA provides a first drop down in the voltage of VBL_TARGET. The target voltage for PBLC is provided at node 2313 below the diode connected PMOS 2311. A resistor R1 2315 provides an additional voltage drop for providing the target voltage for PBLX at node 2317, with a second additional voltage drop provided by resistor R2 2319 for the target voltage of PXXL at node 2321.

FIG. 24 is a diagram of an embodiment of a bias-voltage generation circuit that can provide the bias level used in discharging the sensing node SEN of the sense amplifier embodiments in FIGS. 15 and 22. More specifically, the circuit of FIG. 24 provides the VLOP level below the transistor 1565 of FIGS. 15 and 22 to level that compensates for the threshold voltage Vth of the NMOS transistor 1565 during the discharging of the SEN node in the stabilizing of the sense amplifier in phase 2 as represented in FIGS. 19 and 22.

The input to the circuit of FIG. 24 is the same CLKSA value as applied to the lower plate of the sensing capacitor Csen 1523. During phase 2, when the SEN node and top plate of the are discharged, CLKSA is at its high level of around 1V, for example, before being dropped to its low value of VSSSA at phase 3. To discharge SEN to VLOP+Vth=˜1V, VLOP is biased to 1V-Vth. Connected between the CLKSA input and VSSSA, FIG. 24 includes series connected NMOS transistor 2401 and NMOS transistor 2403. The transistor 2401 can receive a high sense amplifier voltage level VDDSA at its gate so that it is on and the voltage across the transistor corresponds to any drop across BLQ 1571 and STB 1567 when these are on, as in phase 2. NMOS 2403 is diode connected across NMOS 2401, having its gate connected to the CLKSA input. As this mimics the connection of NMOS 1565 to the SEN node, by sizing NMOS 2403 similarly to NMOS 1565 so that they have the same Vth, this can provide the wanted Vth offset from CLKSA in the VLOP. The target VLOP level can be provided from node 2405 between NMOS 2403 and the current source 2407. When CLKSA is subsequently stepped down to 0V in phase 3, this will bring the voltage level on SEN also to ˜0V.

FIG. 25 is a flowchart describing one embodiment of a process for a sensing operation for the sense amplifier of FIG. 15 in the first sensing mode, in which current flows into the sense amplifier. The flow of FIG. 25 corresponds to the phase 1 of FIG. 18, phase 2 of FIG. 19 or alternate embodiment FIG. 22, phase 3 of FIG. 20, and phase 4 of FIG. 21.

Steps 2501 and 2503 are part of the pre-charge phase, or phase 1, as illustrated in FIG. 18. At step 2501 the selected bit line BL 1505, including the selected NAND string or memory cell 1501, and source line CELSRC 1503 are pre-charged to the target level of for CELSRC, where the bit line BL 1505 can be pre-charged through BIAS 1504 from the BLBIAS line. The target value for the voltage level on CELSRC can be provided from the circuit of FIG. 23, which can be part of the control block 1531. At step 2503 the SEN node and top plate of the sensing capacitor Csen 1523 are pre-charged from VHLB though LPC 1573 and BLQ 1571. Steps 2501 and 2503 can be performed concurrently or sequentially in either order, depending on the embodiment.

Steps 2505 and 2507 are part of the settling or stabilization phase, or phase 2, as illustrated in FIG. 19 or FIG. 22. In step 2505 the switch BIAS 1504 is turned off while the source line CELSRC 1503 is maintained at the target level for CELSRC, so that BL 1505 is allowed to settle by discharging through the bit line select switch BLS 1506 and through PBLC 1514 to the low level of SRCGND. In the embodiment of FIG. 19, INV_S is high and PBLC 1514 is cascaded with PBLX 1542 and the path continues through 1561 to SRCGND. In the embodiment of FIG. 22, INV_S can be either high or low and PBLC 1514 is cascaded with PNLO 15419 to SRCGND. At step 2507, the SEN node is discharged though BLQ 1571, STB 1567, and 1565 to VLOP while CLKSA is at −1V. The value on VLOP can be set to a target level by the circuit of FIG. 24, which can be part of the control block 1531. As described above, this allows for local variations of the threshold voltage Vth of NMOS 1565 from sense amplifier to sense amplifier across the memory die to be cancelled. Steps 2505 and 2507 can be performed concurrently or sequentially in either order, depending on the embodiment.

Once the sense amplifier has been pre-charged and settled in phases 1 and 2, in phase 3 the selected memory cell 1501 and elements of the sense amplifier are biased to determine the data state stored on the selected memory cell 1501 as illustrated in FIG. 20. At step 2509 the CLKSA level is stepped down from ˜1V to VSSSA (i.e., ground or 0V). As illustrated in embodiment of FIG. 24, the level on VLOP is based on CLKSA so that when CLKSA goes to ground, so will VLOP. At step 2511, the selected memory cell 1501 is biased to conduct current into the sense amplifier at a rate dependent on the data stored in the memory cell. Biasing the selected memory cell 1501 can include maintaining CELSRC 1503 at its target level and applying a sensing voltage to the word line WL 1502. This process can be used for both verify and data read operations and sensing voltage will depend on the operation and, in multi-level cell (MLC) embodiments, the state being sensed. Depending on the embodiment, the biasing of a selected memory cell can also include applying other bias levels. For example, in the case of a NAND memory embodiment, the non-selected memory cells and select gates of the NAND string of the selected memory cell 1501 will be biased to be on.

The sense amplifier is also biased to accumulate charge on the SEN node and top plate of sensing capacitor Csen 1523 at step 2513, where the amount of charge stored on the SEN node and sensing capacitor Csen 1523 will depend on the data state of the selected memory cell 1501, such as its threshold voltage relative to the sensing voltage applied to the word line WL 1502. In addition to stepping down CLKSA in step 2509 to allow charge to accumulate on the SEN node, PBLC 1514 and PXXL 1522 are turned on, where the voltage levels for the control signals of these devices can be the target values provided from the biasing circuit of FIG. 23. The cascaded PMOSs of PBLC 1514 and PXXL 1522 operate as source followers and allow SEN to charge up based on the current from the selected memory cell 1501 at step 2515 based on the biasing of steps 2511 and 2513.

Along with accumulating charge on the SEN node, the SBUS line is also pre-charged in phase 3 at step 2517. By turning LPC 1573 on, and BLQ 1571 and STB 1567 off, SBUS can be pre-charged from VHLB.

Steps 2519, 2521, and 2523 are part of the strobe phase, or phase 4, as illustrated in FIG. 21. After accumulating the current through the selected memory cell 1501 for an interval in phase 3, at step 2519 the path from BL 1505 to SEN is cut off, such as by turning off PXXL 1522. The level on SEN will determine the amount of current NMOS 1565 will conduct. At step 2521, a strobe is applied to the control gate of STB 1567 so that SBUS will discharge by an amount dependent on the amount of current through NMOS 1565, which is in turn based on the level on SEN, and interval of the duration of the strobe pulse. After the strobe at step 2521, the sensing result is latched into latch 1525.

According to a first set of aspects, a non-volatile memory circuit includes a plurality of memory cells, a bit line connected to one or more of the memory cells, a sense amplifier connectable to the bit line, and one or more control circuits connected to the memory cells and the sense amplifier. The sense amplifier includes a sensing capacitor and a data latch. The one or more control circuits are configured, in a first sensing mode, to: bias a selected one of the memory cells to conduct current into the sense amplifier at a level dependent on a data state stored in the selected memory cell; bias the sense amplifier accumulate charge on the sensing capacitor to an amount dependent on the level of current conducted into the sense amplifier; and latch a first sensing result into the data latch based on the amount of change accumulated on the sensing capacitor.

Other aspects include a method that includes: biasing a selected memory cell to conduct current into a sense amplifier at a level dependent on a data state stored in the selected memory cell; biasing the sense amplifier accumulate charge on a sensing capacitor of the sense amplifier to an amount dependent on the level of current conducted into the sense amplifier; and latching a sensing result into a data latch based on the amount of charge accumulated on the sensing capacitor.

Yet more aspects include a system that includes a sense amplifier having: a first node connectable to a selected memory cell; a sensing capacitor; a sense node connected to the sensing capacitor; a first path between the first node and the sense node; a second path between the first node and sense node; and one or more biasing circuits. The one or more biasing circuits are configured to: sense the selected memory cell in a first sensing mode by charging the sensing capacitor from current conducted along the first path from the selected memory cell; and sense the selected memory cell in a second sensing mode by discharging the sensing capacitor by current conducted along the second path and through the selected memory cell.

For purposes of this document, reference in the specification to “an embodiment,” “one embodiment,” “some embodiments,” or “another embodiment” may be used to describe different embodiments or the same embodiment.

For purposes of this document, a connection may be a direct connection or an indirect connection (e.g., via one or more other parts). In some cases, when an element is referred to as being connected or coupled to another element, the element may be directly connected to the other element or indirectly connected to the other element via intervening elements. When an element is referred to as being directly connected to another element, then there are no intervening elements between the element and the other element. Two devices are “in communication” if they are directly or indirectly connected so that they can communicate electronic signals between them.

For purposes of this document, the term “based on” may be read as “based at least in part on.”

For purposes of this document, without additional context, use of numerical terms such as a “first” object, a “second” object, and a “third” object may not imply an ordering of objects, but may instead be used for identification purposes to identify different objects.

For purposes of this document, the term “set” of objects may refer to a “set” of one or more of the objects.

The foregoing detailed description has been presented for purposes of illustration and description. It is not intended to be exhaustive or to limit to the precise form disclosed. Many modifications and variations are possible in light of the above teaching. The described embodiments were chosen in order to best explain the principles of the proposed technology and its practical application, to thereby enable others skilled in the art to best utilize it in various embodiments and with various modifications as are suited to the particular use contemplated. It is intended that the scope be defined by the claims appended hereto. 

What is claimed is:
 1. A non-volatile memory circuit, comprising: a plurality of memory cells; a bit line connected to one or more of the memory cells; a sense amplifier connectable to the bit line, comprising: a sensing capacitor; and a data latch; and one or more control circuits connected to the memory cells and the sense amplifier and configured, in a first sensing mode, to: bias a selected one of the memory cells to conduct current into the sense amplifier at a level dependent on a data state stored in the selected memory cell; bias the sense amplifier to accumulate charge on the sensing capacitor to an amount dependent on the level of current conducted into the sense amplifier; and latch a first sensing result into the data latch based on the amount of change accumulated on the sensing capacitor.
 2. The non-volatile memory circuit of claim 1, the one or more control circuits further configured, in a second sensing mode, to: pre-charge the sensing capacitor; bias the selected memory cell and the sense amplifier to discharge the sensing capacitor to an amount of charge dependent on the data state stored in the selected memory cell; and latch a second sensing result into the data latch based on the amount of change on the sensing capacitor.
 3. The non-volatile memory circuit of claim 2, wherein the sense amplifier further comprises: a first current path between bit line of the selected memory cell; and a second current path, in parallel with the first current path, between the bit line of the selected memory cell, wherein the one or more control circuits are further configured to, in the first sensing mode, bias the sense amplifier to accumulate charge on the sensing capacitor through the first current path and, in the second sensing mode, discharge the sensing capacitor through the second current path.
 4. The non-volatile memory circuit of claim 3, wherein: the first current path includes a plurality of PMOS transistors connected in series between the bit line of the sensing capacitor; and the second current path includes a plurality of NMOS transistors connected in series between the bit line of the sensing capacitor.
 5. The non-volatile memory circuit of claim 1, wherein: the data state stored in the selected memory cell is a negative threshold data state.
 6. The non-volatile memory circuit of claim 1, wherein: the one or more memory cells, including the selected memory cell, are part of a NAND string; and the one or more control circuits are configured to: concurrently with biasing the selected memory cell to conduct current into the sense amplifier at a level dependent on a data state stored in the selected memory cell, bias non-selected memory cells of the NAND string and select gates of the NAND string to a conducting state.
 7. A method, comprising: biasing a selected memory cell to conduct current into a sense amplifier at a level dependent on a data state stored in the selected memory cell; biasing the sense amplifier to accumulate charge on a sensing capacitor of the sense amplifier to an amount dependent on the level of current conducted into the sense amplifier; and latching a sensing result into a data latch based on the amount of charge accumulated on the sensing capacitor.
 8. The method of claim 7, wherein biasing the sense amplifier accumulate charge on a sensing capacitor of the sense amplifier includes: biasing a pair of PMOS transistors connected in series between a bit line to which the selected memory cell is connected and the sensing capacitor to be in an on state.
 9. The method of claim 7, wherein latching a sensing result into a data latch based on the amount of charge accumulated on the sensing capacitor includes: pre-charging an internal bus line connected to a data latch; discharging for a sensing interval the internal bus line through a sensing transistor having a control gate connected to the sensing capacitor; and subsequently latching the sensing result in the data latch based a voltage level on the internal bus line.
 10. The method of claim 9, wherein biasing the sense amplifier accumulate charge on a sensing capacitor of the sense amplifier includes: prior to accumulating charge on the sensing capacitor, pre-charging the sensing capacitor; and discharging the sensing capacitor to a level to offset a threshold voltage of the sensing transistor.
 11. The method of claim 7, wherein biasing the sense amplifier to accumulate charge on a sensing capacitor of the sense amplifier includes: prior to accumulating charge on the sensing capacitor, pre-charging a bit line to which the selected memory cell is connected.
 12. The method of claim 11, wherein biasing the sense amplifier accumulate charge on a sensing capacitor of the sense amplifier includes: prior to accumulating charge on the sensing capacitor and subsequent to pre-charging the bit line to which the selected memory cell is connected, discharging the bit line to which the selected memory cell is connected through a pair of series connected PMOS devices.
 13. A sense amplifier comprising: a first node connectable to a selected memory cell; a sensing capacitor; a sense node connected to the sensing capacitor; a first path between the first node and the sense node; a second path between the first node and sense node; and one or more biasing circuits configured to: sense the selected memory cell in a first sensing mode by charging the sensing capacitor from current conducted along the first path from the selected memory cell; and sense the selected memory cell in a second sensing mode by discharging the sensing capacitor by current conducted along the second path and through the selected memory cell.
 14. The sense amplifier of claim 13, wherein: the first path includes a first PMOS transistor connected in series with a second PMOS transistor between the first node and the sense node; and the second path includes a first NMOS transistor connected in series with a second NMOS transistor between the first node and the sense node, wherein the first PMOS transistor is connected in parallel with the first NMOS transistor and the second PMOS transistor is connected in parallel with the second NMOS.
 15. The sense amplifier of claim 14, further comprising: a third PMOS transistor, wherein the one or more biasing circuits are further configured to: connect the third PMOS transistor in series with the first PMOS transistor in the first sensing mode to discharge first node to a low voltage level.
 16. The sense amplifier of claim 15, wherein the one or more biasing circuits are further configured to: prior to discharging first node to a low voltage level, pre-charging the first node.
 17. The sense amplifier of claim 13, further comprising: a bit line select switch whereby the first node is selectively connectable to a plurality of bit lines including a bit line to which the selected memory cell is connected.
 18. The sense amplifier of claim 13, further comprising: a data latch; an internal sensing bus line connected to the data latch; and a discharge transistor having a control gate connected to the sense node, wherein the one or more biasing circuits are further configured to: subsequent to charging the sensing capacitor in the first sensing mode, discharging the internal sensing bus line through the discharge transistor for a sensing interval; and subsequent to discharging the internal sensing bus line for the sensing interval, latch a sensing result based on a voltage level on the internal sensing bus line into the data latch.
 19. The sense amplifier of claim 18, wherein the one or more biasing circuits are further configured to: prior to discharging the internal sensing bus line for the sensing interval, pre-charging the internal sensing bus line.
 20. The sense amplifier of claim 18, wherein the one or more biasing circuits are further configured to: prior to charging the sensing capacitor in the first sensing mode, pre-charge the sensing capacitor and subsequently discharge the sensing capacitor to level to compensate for a threshold voltage of the discharge transistor. 